Method for estimating CFO in wireless LAN system that uses 16 QAM

ABSTRACT

Disclosed is a method for estimating a CFO, the method comprising: measuring a size ratio of a signal pair received in two consecutive OFDM symbols of a predetermined subcarrier; determining the received signal pair as a signal pair to be used for estimating a residual CFO when the size ratio is within a first threshold range; and estimating the residual CFO, using only the received signal pair which has been determined to be used for estimating the residual CFO.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is the National Stage filing under 35 U.S.C. 371 of International Application No. PCT/KR2016/000594, filed on Jan. 20, 2016, which claims the benefit of U.S. Provisional Application No. 62/172,772, filed on Jun. 8, 2015, the contents of which are all hereby incorporated by reference herein in their entirety.

TECHNICAL FIELD

Following description relates to a wireless communication system, and more particularly, to a method of estimating a CFO in a wireless LAN system using 16 QAM and an apparatus therefor.

BACKGROUND ART

Recently, with development of information communication technology, various wireless communication technologies have been developed. Among others, a wireless local area network (WLAN) enables wireless access to the Internet using a portable terminal such as a personal digital assistant (PDA), a laptop, a portable multimedia player (PMP) in a home, an enterprise or a specific service provision area based on radio frequency technology.

In order to overcome limitations in communication rate which have been pointed out as weakness of a WLAN, in recent technical standards, a system for increasing network speed and reliability and extending wireless network distance has been introduced. For example, in IEEE 802.11n, multiple input and multiple output (MIMO) technology using multiple antennas in a transmitter and a receiver has been introduced in order to support high throughput (HT) with a maximum data rate of 540 Mbps or more, to minimize transmission errors, and to optimize data rate.

As next-generation communication technology, machine-to-machine (M2M) communication technology has been discussed. Even in an IEEE 802.11 WLAN system, technical standards supporting M2M communication have been developed as IEEE 802.11ah. In M2M communication, a scenario in which a small amount of data is communicated at a low rate may be considered in an environment in which many apparatuses are present.

Communication in a WLAN system is performed in a medium shared between all apparatuses. As in M2M communication, if the number of apparatuses is increased, in order to reduce unnecessary power consumption and interference, a channel access mechanism needs to be more efficiently improved.

DISCLOSURE OF THE INVENTION Technical Tasks

The present invention is devised to solve the above-mentioned problems and an object of the present invention is to enable a reception module to estimate a CFO accurately.

Another object of the present invention is to increase accuracy of estimating a CFO in a manner that a reception module sufficiently secures the number of samples to be utilized for estimating the CFO.

The other object of the present invention is to apply a CFO estimation method of a blind type to a high order modulation scheme.

The technical problems solved by the present invention are not limited to the above technical problems and other technical problems which are not described herein will become apparent to those skilled in the art from the following description.

Technical Solution

To achieve these and other advantages and in accordance with the purpose of the present invention, as embodied and broadly described, according to one embodiment, a method of estimating a CFO (carrier frequency offset), which is estimated by a reception module in a wireless communication system using 16-QAM (quadrature amplitude modulation), includes the steps of measuring a size ratio between a pair of received signals received on two contiguous (or consecutive) OFDM (orthogonal frequency division multiplexing) symbols for a specific subcarrier, if the size ratio belongs to a first threshold range, determining the pair of received signals as a pair to be used for estimating a residual CFO, and estimating the residual CFO using only pairs of received signals which are determined to be used for estimating the residual CFO.

The method can further include the step of measuring a phase difference between the pair of received signals. In this case, if the size ratio belongs to the first threshold range and the phase difference belongs to a second threshold range, the determining step can determine the pair of received signals as a pair to be used for estimating the residual CFO.

If the phase difference does not belong to the second threshold range and belongs to a third threshold range, the method can further include the step of changing a phase of multiplication of the pair of received signals as much as a predetermined value.

If the phase difference does not belong to the second threshold range and does not belong to the third threshold range, the method can further include the step of determining not to use the pair of received signals for estimating the residual CFO.

The second threshold range may indicate a range within a predetermined value from one value among 0°, 90°, 180°, and 270°.

The third threshold range may indicate a range within a predetermined value from one value among 46°, 136°, 226°, and 316°.

The first threshold range may indicate a range within a predetermined value from one value among 1, ⅓, and 3.

The step of estimating the residual CFO can estimate the residual CFO according to a blind scheme which is applied when QPSK (quadrature phase shift keying) is used.

To further achieve these and other advantages and in accordance with the purpose of the present invention, according to a different embodiment, a reception module estimating a CFO in a wireless communication system using 16-QAM (quadrature amplitude modulation) includes a transmitter, a receiver, and a processor configured to operate in a manner of being connected with the transmitter and the receiver, the processor configured to measure a size ratio between a pair of received signals received on two contiguous OFDM symbols for a specific subcarrier, the processor, if the size ratio belongs to a first threshold range, configured to determine the pair of received signals as a pair to be used for estimating a residual CFO, the processor configured to estimate the residual CFO using only pairs of received signals which are determined to be used for estimating the residual CFO.

Advantageous Effects

According to the embodiments of the present invention, it is able to have the following effects.

First of all, it is able to increase a decoding efficiency of a reception signal in a manner that a reception module precisely measures a residual CFO.

Second, if the number of samples utilized for estimating a CFO is increased, it is able to more efficiently perform a blind CFO estimation scheme.

Third, it is able to estimate a CFO by utilizing a blind type even in 16 QAM corresponding to a high order compared to BPSK, QPSK, and the like.

The effects of the present invention are not limited to the above-described effects and other effects which are not described herein may be derived by those skilled in the art from the following description of the embodiments of the present invention. That is, effects which are not intended by the present invention may be derived by those skilled in the art from the embodiments of the present invention.

DESCRIPTION OF DRAWINGS

The accompanying drawings, which are included to provide a further understanding of the invention, illustrate embodiments of the invention and together with the description serve to explain the principle of the invention. The technical features of the present invention are not limited to specific drawings and the features shown in the drawings are combined to construct a new embodiment. Reference numerals of the drawings mean structural elements.

FIG. 1 is a diagram showing an exemplary structure of an IEEE 802.11 system to which the present invention is applicable;

FIG. 2 is a diagram showing another exemplary structure of an IEEE 802.11 system to which the present invention is applicable;

FIG. 3 is a diagram showing another exemplary structure of an IEEE 802.11 system to which the present invention is applicable;

FIG. 4 is a diagram showing an exemplary structure of a WLAN system;

FIG. 5 is a diagram illustrating a link setup process in a WLAN system;

FIG. 6 is a diagram illustrating a backoff process;

FIG. 7 is a diagram illustrating a hidden node and an exposed node;

FIG. 8 is a diagram illustrating request to send (RTS) and clear to send (CTS);

FIG. 9 is a diagram illustrating power management operation;

FIGS. 10 to 12 are diagrams illustrating operation of a station (STA) which receives a traffic indication map (TIM);

FIG. 13 is a diagram illustrating a group based association identifier (AID);

FIGS. 14 to 16 are diagrams showing examples of operation of an STA if a group channel access interval is set;

FIGS. 17 and 19 are diagrams illustrating frame structures according to the present invention and constellations thereof;

FIG. 20 is a diagram illustrating frequency-domain pilot signals according to the present invention;

FIGS. 21 and 22 are diagrams for explaining a method of estimating a CFO according to the present invention;

FIG. 23 is a flowchart for a method of estimating a CFO according to the present invention;

FIGS. 24 to 26 are diagrams for explaining a method of estimating a CFO according to the present invention;

FIG. 27 is a flowchart for a method of estimating a CFO according to the present invention;

FIG. 28 is a diagram illustrating a resource block according to the present invention;

FIG. 29 is a flowchart for a method of estimating a CFO according to the present invention;

FIGS. 30 and 31 are diagrams for a method of segmenting 16-QAM constellation according to embodiments proposed in the present invention;

FIG. 32 is a flowchart for a method of estimating a CFO according to the present invention;

FIG. 33 is a diagram for configurations of a user equipment and a base station according to one embodiment.

BEST MODE Mode for Invention

Although the terms used in the present invention are selected from generally known and used terms, terms used herein may be varied depending on operator's intention or customs in the art, appearance of new technology, or the like. In addition, some of the terms mentioned in the description of the present invention have been selected by the applicant at his or her discretion, the detailed meanings of which are described in relevant parts of the description herein. Furthermore, it is required that the present invention is understood, not simply by the actual terms used but by the meanings of each term lying within.

The following embodiments are proposed by combining constituent components and characteristics of the present invention according to a predetermined format. The individual constituent components or characteristics should be considered optional factors on the condition that there is no additional remark. If required, the individual constituent components or characteristics may not be combined with other components or characteristics. In addition, some constituent components and/or characteristics may be combined to implement the embodiments of the present invention. The order of operations to be disclosed in the embodiments of the present invention may be changed. Some components or characteristics of any embodiment may also be included in other embodiments, or may be replaced with those of the other embodiments as necessary.

In describing the present invention, if it is determined that the detailed description of a related known function or construction renders the scope of the present invention unnecessarily ambiguous, the detailed description thereof will be omitted.

In the entire specification, when a certain portion “comprises or includes” a certain component, this indicates that the other components are not excluded and may be further included unless specially described otherwise. The terms “unit”, “-or/er” and “module” described in the specification indicate a unit for processing at least one function or operation, which may be implemented by hardware, software or a combination thereof. The words “a or an”, “one”, “the” and words related thereto may be used to include both a singular expression and a plural expression unless the context describing the present invention (particularly, the context of the following claims) clearly indicates otherwise.

In this document, the embodiments of the present invention have been described centering on a data transmission and reception relationship between a mobile station and a base station. The base station may mean a terminal node of a network which directly performs communication with a mobile station. In this document, a specific operation described as performed by the base station may be performed by an upper node of the base station.

Namely, it is apparent that, in a network comprised of a plurality of network nodes including a base station, various operations performed for communication with a mobile station may be performed by the base station, or network nodes other than the base station. The term base station may be replaced with the terms fixed station, Node B, eNode B (eNB), advanced base station (ABS), access point, etc.

The term mobile station (MS) may be replaced with user equipment (UE), subscriber station (SS), mobile subscriber station (MSS), mobile terminal, advanced mobile station (AMS), terminal, etc.

A transmitter refers to a fixed and/or mobile node for transmitting a data or voice service and a receiver refers to a fixed and/or mobile node for receiving a data or voice service. Accordingly, in uplink, a mobile station becomes a transmitter and a base station becomes a receiver. Similarly, in downlink transmission, a mobile station becomes a receiver and a base station becomes a transmitter.

Communication of a device with a “cell” may mean that the device transmit and receive a signal to and from a base station of the cell. That is, although a device substantially transmits and receives a signal to a specific base station, for convenience of description, an expression “transmission and reception of a signal to and from a cell formed by the specific base station” may be used. Similarly, the term “macro cell” and/or “small cell” may mean not only specific coverage but also a “macro base station supporting the macro cell” and/or a “small cell base station supporting the small cell”.

The embodiments of the present invention can be supported by the standard documents disclosed in any one of wireless access systems, such as an IEEE 802.xx system, a 3rd Generation Partnership Project (3GPP) system, a 3GPP Long Term Evolution (LTE) system, and a 3GPP2 system. That is, the steps or portions, which are not described in order to make the technical spirit of the present invention clear, may be supported by the above documents.

In addition, all the terms disclosed in the present document may be described by the above standard documents. In particular, the embodiments of the present invention may be supported by at least one of P802.16-2004, P802.16e-2005, P802.16.1, P802.16p and P802.16.1b documents, which are the standard documents of the IEEE 802.16 system.

Hereinafter, the preferred embodiments of the present invention will be described with reference to the accompanying drawings. It is to be understood that the detailed description which will be disclosed along with the accompanying drawings is intended to describe the exemplary embodiments of the present invention, and is not intended to describe a unique embodiment which the present invention can be carried out.

It should be noted that specific terms disclosed in the present invention are proposed for convenience of description and better understanding of the present invention, and the use of these specific terms may be changed to another format within the technical scope or spirit of the present invention.

1. IEEE 802.11 System Overview

1.1 Structure of WLAN System

FIG. 1 is a diagram showing an exemplary structure of an IEEE 802.11 system to which the present invention is applicable.

An IEEE 802.11 structure may be composed of a plurality of components and a wireless local area network (WLAN) supporting station (STA) mobility transparent to a higher layer may be provided by interaction among the components. A basic service set (BSS) may correspond to a basic component block in an IEEE 802.11 LAN. In FIG. 1, two BSSs (BSS1 and BSS2) are present and each BSS includes two STAs (STA1 and STA2 are included in BSS1 and STA3 and STA4 are included in BSS2) as members. In FIG. 1, an ellipse indicating the BSS indicates a coverage area in which STAs included in the BSS maintains communication. This area may be referred to as a basic service area (BSA). If an STA moves out of a BSA, the STA cannot directly communicate with other STAs in the BSA.

In an IEEE 802.11 LAN, a BSS is basically an independent BSS (IBSS). For example, the IBSS may have only two STAs. In addition, the simplest BSS (BSS1 or BSS2) of FIG. 1, in which other components are omitted, may correspond to a representative example of the IBSS. Such a configuration is possible when STAs can directly perform communication. In addition, such a LAN is not configured in advance but may be configured if a LAN is necessary. This LAN may also be referred to as an ad-hoc network.

If an STA is turned on or off or if an STA enters or moves out of a BSS, the membership of the STA in the BSS may be dynamically changed. An STA may join a BSS using a synchronization process in order to become a member of the BSS. In order to access all services of a BSS based structure, an STA should be associated with the BSS. Such association may be dynamically set and may include use of a distribution system service (DSS).

FIG. 2 is a diagram showing another exemplary structure of an IEEE 802.11 system to which the present invention is applicable. In FIG. 2, a distribution system (DS), a distribution system medium (DSM) and an access point (AP) are added to the structure of FIG. 1.

In a LAN, a direct station-to-station distance may be restricted by PHY performance Although such distance restriction may be possible, communication between stations located at a longer distance may be necessary. In order to support extended coverage, a DS may be configured.

The DS means a structure in which BSSs are mutually connected. More specifically, the BSSs are not independently present as shown in FIG. 1 but the BSS may be present as an extended component of a network including a plurality of BSSs.

The DS is a logical concept and may be specified by characteristics of the DSM. In IEEE 802.11 standards, a wireless medium (WM) and a DSM are logically distinguished. Logical media are used for different purposes and are used by different components. In IEEE 802.11 standards, such media are not restricted to the same or different media. Since plural media are logically different, an IEEE 802.11 LAN structure (a DS structure or another network structure) may be flexible. That is, the IEEE 802.11 LAN structure may be variously implemented and a LAN structure may be independently specified by physical properties of each implementation.

The DS provides seamless integration of a plurality of BSSs and provides logical services necessary to treat an address to a destination so as to support a mobile apparatus.

The AP means an entity which enables associated STAs to access the DS via the WM and has STA functionality. Data transfer between the BSS and the DS may be performed via the AP. For example, STA2 and STA3 shown in FIG. 2 have STA functionality and provide a function enabling associated STAs (STA1 and STA4) to access the DS. In addition, since all APs correspond to STAs, all APs may be addressable entities. An address used by the AP for communication on the WM and an address used by the AP for communication on the DSM may not be equal.

Data transmitted from one of STAs associated with the AP to the STA address of the AP may always be received by an uncontrolled port and processed by an IEEE 802.1X port access entity. In addition, if a controlled port is authenticated, transmission data (or frames) may be transmitted to the DS.

FIG. 3 is a diagram showing another exemplary structure of an IEEE 802.11 system to which the present invention is applicable. In FIG. 3, an extended service set (ESS) for providing wide coverage is added to the structure of FIG. 2.

A wireless network having an arbitrary size and complexity may be composed of a DS and BSSs. In an IEEE 802.11 system, such a network is referred to as an ESS network. The ESS may correspond to a set of BSSs connected to one DS. However, the ESS does not include the DS. The ESS network appears as an IBSS network at a logical link control (LLC) layer. STAs included in the ESS may communicate with each other and mobile STAs may move from one BSS to another BSS (within the same ESS) transparently to the LLC layer.

In IEEE 802.11, relative physical locations of the BSSs in FIG. 3 are not assumed and may be defined as follows. The BSSs may partially overlap in order to provide consecutive coverage. In addition, the BSSs may not be physically connected and a distance between BSSs is not logically restricted. In addition, the BSSs may be physically located at the same location in order to provide redundancy. In addition, one (or more) IBSS or ESS network may be physically present in the same space as one (or more) ESS network. This corresponds to an ESS network type such as a case in which an ad-hoc network operates at a location where the ESS network is present, a case in which IEEE 802.11 networks physically overlapped by different organizations are configured or a case in which two or more different access and security policies are necessary at the same location.

FIG. 4 is a diagram showing an exemplary structure of a WLAN system. FIG. 4 shows an example of an infrastructure BSS including a DS.

In the example of FIG. 4, BSS1 and BSS2 configure an ESS. In the WLAN system, an STA operates according to a MAC/PHY rule of IEEE 802.11. The STA includes an AP STA and a non-AP STA. The non-AP STA corresponds to an apparatus directly handled by a user, such as a laptop or a mobile phone. In the example of FIG. 4, STA1, STA3 and STA4 correspond to the non-AP STA and STA2 and STA5 correspond to the AP STA.

In the following description, the non-AP STA may be referred to as a terminal, a wireless transmit/receive unit (WTRU), a user equipment (UE), a mobile station (MS), a mobile terminal or a mobile subscriber station (MSS). In addition, the AP may correspond to a base station (BS), a Node-B, an evolved Node-B (eNB), a base transceiver system (BTS) or a femto BS.

1.2 Link Setup Process

FIG. 5 is a diagram illustrating a general link setup process.

In order to establish a link with respect to a network and perform data transmission and reception, an STA discovers the network, performs authentication, establishes association and performs an authentication process for security. The link setup process may be referred to as a session initiation process or a session setup process. In addition, discovery, authentication, association and security setup of the link setup process may be collectively referred to as an association process.

An exemplary link setup process will be described with reference to FIG. 5.

In step S510, the STA may perform a network discovery operation. The network discovery operation may include a scanning operation of the STA. That is, the STA discovers the network in order to access the network. The STA should identify a compatible network before participating in a wireless network and a process of identifying a network present in a specific area is referred to as scanning. The scanning method includes an active scanning method and a passive scanning method.

In FIG. 5, a network discovery operation including an active scanning process is shown. In active scanning, the STA which performs scanning transmits a probe request frame while moving between channels and waits for a response thereto, in order to detect which AP is present. A responder transmits a probe response frame to the STA, which transmitted the probe request frame, as a response to the probe request frame. The responder may be an STA which lastly transmitted a beacon frame in a BSS of a scanned channel. In the BSS, since the AP transmits the beacon frame, the AP is the responder. In the IBSS, since the STAs in the IBSS alternately transmit the beacon frame, the responder is not fixed. For example, the STA which transmits the probe request frame on a first channel and receives the probe response frame on the first channel stores BSS related information included in the received probe response frame, moves to a next channel (e.g., a second channel) and performs scanning (probe request/response transmission/reception on the second channel) using the same method.

Although not shown in FIG. 5, a scanning operation may be performed using a passive scanning method. In passive scanning, the STA which performs scanning waits for a beacon frame while moving between channels. The beacon frame is a management frame in IEEE 802.11 and is periodically transmitted in order to indicate presence of a wireless network and to enable the STA, which performs scanning, to discover and participate in the wireless network. In the BSS, the AP is responsible for periodically transmitting the beacon frame. In the IBSS, the STAs alternately transmit the beacon frame. The STA which performs scanning receives the beacon frame, stores information about the BSS included in the beacon frame, and records beacon frame information of each channel while moving to another channel. The STA which receives the beacon frame may store BSS related information included in the received beacon frame, move to a next channel and perform scanning on the next channel using the same method.

Active scanning has delay and power consumption less than those of passive scanning.

After the STA has discovered the network, an authentication process may be performed in step S520. Such an authentication process may be referred to as a first authentication process to be distinguished from a security setup operation of step S540.

The authentication process includes a process of, at the STA, transmitting an authentication request frame to the AP and, at the AP, transmitting an authentication response frame to the STA in response thereto. The authentication frame used for authentication request/response corresponds to a management frame.

The authentication frame may include information about an authentication algorithm number, an authentication transaction sequence number, a status code, a challenge text, a robust security network (RSN), a finite cyclic group, etc. The information may be examples of information included in the authentication request/response frame and may be replaced with other information. The information may further include additional information.

The STA may transmit the authentication request frame to the AP. The AP may determine whether authentication of the STA is allowed, based on the information included in the received authentication request frame. The AP may provide the STA with the authentication result via the authentication response frame.

After the STA is successfully authenticated, an association process may be performed in step S530. The association process includes a process of, at the STA, transmitting an association request frame to the AP and, at the AP, transmitting an association response frame to the STA in response thereto.

For example, the association request frame may include information about various capabilities, beacon listen interval, service set identifier (SSID), supported rates, RSN, mobility domain, supported operating classes, traffic indication map (TIM) broadcast request, interworking service capability, etc.

For example, the association response frame may include information about various capabilities, status code, association ID (AID), supported rates, enhanced distributed channel access (EDCA) parameter set, received channel power indicator (RCPI), received signal to noise indicator (RSNI), mobility domain, timeout interval (association comeback time), overlapping BSS scan parameter, TIM broadcast response, QoS map, etc.

This information is purely exemplary information included in the association request/response frame and may be replaced with other information. This information may further include additional information.

After the STA is successfully authenticated, a security setup process may be performed in step S540. The security setup process of step S540 may be referred to as an authentication process through a robust security network association (RSNA) request/response. The authentication process of step S520 may be referred to as the first authentication process and the security setup process of step S540 may be simply referred to as an authentication process.

The security setup process of step S540 may include a private key setup process through 4-way handshaking of an extensible authentication protocol over LAN (EAPOL) frame. In addition, the security setup process may be performed according to a security method which is not defined in the IEEE 802.11 standard.

2.1 Evolution of WLAN

As a technical standard recently established in order to overcome limitations in communication speed in a WLAN, IEEE 802.11n has been devised. IEEE 802.11n aims at increasing network speed and reliability and extending wireless network distance. More specifically, IEEE 802.11n is based on multiple input and multiple output (MIMO) technology using multiple antennas in a transmitter and a receiver in order to support high throughput (HT) with a maximum data rate of 540 Mbps or more, to minimize transmission errors, and to optimize data rate.

As WLANs have come into widespread use and applications using the same have been diversified, recently, there is a need for a new WLAN system supporting throughput higher than a data rate supported by IEEE 802.11n. A next-generation WLAN system supporting very high throughput (VHT) is a next version (e.g., IEEE 802.11ac) of the IEEE 802.11n WLAN system and is an IEEE 802.11 WLAN system newly proposed in order to support a data rate of 1 Gbps or more at a MAC service access point (SAP).

The next-generation WLAN system supports a multi-user MIMO (MU-MIMO) transmission scheme by which a plurality of STAs simultaneously accesses a channel in order to efficiently use a radio channel. According to the MU-MIMO transmission scheme, the AP may simultaneously transmit packets to one or more MIMO-paired STAs.

In addition, support of a WLAN system operation in a whitespace is being discussed. For example, introduction of a WLAN system in a TV whitespace (WS) such as a frequency band (e.g., 54 to 698 MHz) in an idle state due to digitalization of analog TVs is being discussed as the IEEE 802.11af standard. However, this is only exemplary and the whitespace may be incumbently used by a licensed user. The licensed user means a user who is allowed to use a licensed band and may be referred to as a licensed device, a primary user or an incumbent user.

For example, the AP and/or the STA which operate in the WS should provide a protection function to the licensed user. For example, if a licensed user such as a microphone already uses a specific WS channel which is a frequency band divided on regulation such that a WS band has a specific bandwidth, the AP and/or the STA cannot use the frequency band corresponding to the WS channel in order to protect the licensed user. In addition, the AP and/or the STA must stop use of the frequency band if the licensed user uses the frequency band used for transmission and/or reception of a current frame.

Accordingly, the AP and/or the STA should perform a procedure of determining whether a specific frequency band in a WS band is available, that is, whether a licensed user uses the frequency band. Determining whether a licensed user uses a specific frequency band is referred to as spectrum sensing. As a spectrum sensing mechanism, an energy detection method, a signature detection method, etc. may be used. It may be determined that the licensed user uses the frequency band if received signal strength is equal to or greater than a predetermined value or if a DTV preamble is detected.

In addition, as next-generation communication technology, machine-to-machine (M2M) communication technology is being discussed. Even in an IEEE 802.11 WLAN system, a technical standard supporting M2M communication has been developed as IEEE 802.11ah. M2M communication means a communication scheme including one or more machines and may be referred to as machine type communication (MTC). Here, a machine means an entity which does not require direct operation or intervention of a person. For example, a device including a mobile communication module, such as a meter or a vending machine, may include a user equipment such as a smart phone which is capable of automatically accessing a network without operation/intervention of a user to perform communication. M2M communication includes communication between devices (e.g., device-to-device (D2D) communication) and communication between a device and an application server. Examples of communication between a device and a server include communication between a vending machine and a server, communication between a point of sale (POS) device and a server and communication between an electric meter, a gas meter or a water meter and a server. An M2M communication based application may include security, transportation, health care, etc. If the characteristics of such examples are considered, in general, M2M communication should support transmission and reception of a small amount of data at a low rate in an environment in which very many apparatuses are present.

More specifically, M2M communication should support a larger number of STAs. In a currently defined WLAN system, it is assumed that a maximum of 2007 STAs is associated with one AP. However, in M2M communication, methods supporting the case in which a larger number of STAs (about 6000) are associated with one AP are being discussed. In addition, in M2M communication, it is estimated that there are many applications supporting/requiring a low transfer rate. In order to appropriately support the low transfer rate, for example, in a WLAN system, the STA may recognize presence of data to be transmitted thereto based on a traffic indication map (TIM) element and methods of reducing a bitmap size of the TIM are being discussed. In addition, in M2M communication, it is estimated that there is traffic having a very long transmission/reception interval. For example, in electricity/gas/water consumption, a very small amount of data is required to be exchanged at a long period (e.g., one month). In a WLAN system, although the number of STAs associated with one AP is increased, methods of efficiently supporting the case in which the number of STAs, in which a data frame to be received from the AP is present during one beacon period, is very small are being discussed.

WLAN technology has rapidly evolved. In addition to the above-described examples, technology for direct link setup, improvement of media streaming performance, support of fast and/or large-scale initial session setup, support of extended bandwidth and operating frequency, etc. is being developed.

2.2 Medium Access Mechanism

In a WLAN system according to IEEE 802.11, the basic access mechanism of medium access control (MAC) is a carrier sense multiple access with collision avoidance (CSMA/CA) mechanism. The CSMA/CA mechanism is also referred to as a distributed coordination function (DCF) of IEEE 802.11 MAC and employs a “listen before talk” access mechanism. According to such an access mechanism, the AP and/or the STA may perform clear channel assessment (CCA) for sensing a radio channel or medium during a predetermined time interval (for example, a DCF inter-frame space (DIFS)) before starting transmission. If it is determined that the medium is in an idle state as the sensed result, frame transmission starts via the medium. If it is determined that the medium is in an occupied state, the AP and/or the STA may set and wait for a delay period (e.g., a random backoff period) for medium access without starting transmission and then attempt to perform frame transmission. Since several STAs attempt to perform frame transmission after waiting for different times by applying the random backoff period, it is possible to minimize collision.

In addition, the IEEE 802.11 MAC protocol provides a hybrid coordination function (HCF). The HCF is based on the DCF and a point coordination function (PCF). The PCF refers to a periodic polling method for enabling all reception AP and/or STAs to receive data frames using a polling based synchronous access method. In addition, the HCF has enhanced distributed channel access (EDCA) and HCF controlled channel access (HCCA). The EDCA uses a contention access method for providing data frames to a plurality of users by a provider and the HCCA uses a contention-free channel access method using a polling mechanism. In addition, the HCF includes a medium access mechanism for improving quality of service (QoS) of a WLAN and may transmit QoS data both in a contention period (CP) and a contention free period (CFP).

FIG. 6 is a diagram illustrating a backoff process.

Operation based on a random backoff period will be described with reference to FIG. 6. If a medium is changed from an occupied or busy state to an idle state, several STAs may attempt data (or frame) transmission. At this time, a method of minimizing collision, the STAs may select respective random backoff counts, wait for slot times corresponding to the random backoff counts and attempt transmission. The random backoff count has a pseudo-random integer and may be set to one of values of 0 to CW. Here, the CW is a contention window parameter value. The CW parameter is set to CWmin as an initial value but may be set to twice CWmin if transmission fails (e.g., ACK for the transmission frame is not received). If the CW parameter value becomes CWmax, data transmission may be attempted while maintaining the CWmax value until data transmission is successful. If data transmission is successful, the CW parameter value is reset to CWmin. CW, CWmin and CWmax values are preferably set to 2n−1 (n=0, 1, 2, . . . ).

If the random backoff process starts, the STA continuously monitors the medium while the backoff slots are counted down according to the set backoff count value. If the medium is in the occupied state, countdown is stopped and, if the medium is in the idle state, countdown is resumed.

In the example of FIG. 6, if packets to be transmitted to the MAC of STA3 arrive, STA3 may confirm that the medium is in the idle state during the DIFS and immediately transmit a frame. Meanwhile, the remaining STAs monitor that the medium is in the busy state and wait. During a wait time, data to be transmitted may be generated in STA1, STA2 and STA5. The STAs may wait for the DIFS if the medium is in the idle state and then count down the backoff slots according to the respectively selected random backoff count values.

In the example of FIG. 6, STA2 selects a smallest backoff count value and STA1 selects a largest backoff count value. That is, the residual backoff time of STA5 is less than the residual backoff time of STA1 when STA2 completes backoff count and starts frame transmission. STA1 and STA5 stop countdown and wait while STA2 occupies the medium. If occupancy of the medium by STA2 ends and the medium enters the idle state again, STA1 and STA5 wait for the DIFS and then resume countdown. That is, after the residual backoff slots corresponding to the residual backoff time are counted down, frame transmission may start. Since the residual backoff time of STA5 is less than of STA1, STA5 starts frame transmission.

If STA2 occupies the medium, data to be transmitted may be generated in the STA4. At this time, STA4 may wait for the DIFS if the medium enters the idle state, perform countdown according to a random backoff count value selected thereby, and start frame transmission. In the example of FIG. 6, the residual backoff time of STA5 accidentally matches the random backoff time of STA4. In this case, collision may occur between STA4 and STA5. If collision occurs, both STA4 and STA5 do not receive ACK and data transmission fails. In this case, STA4 and STA5 may double the CW value, select the respective random backoff count values and then perform countdown. STA1 may wait while the medium is busy due to transmission of STA4 and STA5, wait for the DIFS if the medium enters the idle state, and start frame transmission if the residual backoff time has elapsed.

2.3 Sensing Operation of STA

As described above, the CSMA/CA mechanism includes not only physical carrier sensing for directly sensing a medium by an AP and/or an STA but also virtual carrier sensing. Virtual carrier sensing solves a problem which may occur in medium access, such as a hidden node problem. For virtual carrier sensing, MAC of a WLAN may use a network allocation vector (NAV). The NAV refers to a value of a time until a medium becomes available, which is indicated to another AP and/or STA by an AP and/or an STA, which is currently utilizing the medium or has rights to utilize the medium. Accordingly, the NAV value corresponds to a period of time when the medium will be used by the AP and/or the STA for transmitting the frame, and medium access of the STA which receives the NAV value is prohibited during that period of time. The NAV may be set according to the value of the “duration” field of a MAC header of a frame.

A robust collision detection mechanism for reducing collision has been introduced, which will be described with reference to FIGS. 7 and 8. Although a transmission range may not be equal to an actual carrier sensing range, for convenience, assume that the transmission range may be equal to the actual carrier sensing range.

FIG. 7 is a diagram illustrating a hidden node and an exposed node.

FIG. 7(a) shows a hidden node, and, in this case, an STA A and an STA B are performing communication and an STA C has information to be transmitted. More specifically, although the STA A transmits information to the STA B, the STA C may determine that the medium is in the idle state, when carrier sensing is performed before transmitting data to the STA B. This is because the STA C may not sense transmission of the STA A (that is, the medium is busy). In this case, since the STA B simultaneously receives information of the STA A and the STA C, collision occurs. At this time, the STA A may be the hidden node of the STA C.

FIG. 7(b) shows an exposed node and, in this case, the STA B transmits data to the STA A and the STA C has information to be transmitted to the STA D. In this case, if the STA C performs carrier sensing, it may be determined that the medium is busy due to transmission of the STA B. If the STA C has information to be transmitted to the STA D, since it is sensed that the medium is busy, the STA C waits until the medium enters the idle state. However, since the STA A is actually outside the transmission range of the STA C, transmission from the STA C and transmission from the STA B may not collide from the viewpoint of the STA A. Therefore, the STA C unnecessarily waits until transmission of the STA B is stopped. At this time, the STA C may be the exposed node of the STA B.

FIG. 8 is a diagram illustrating request to send (RTS) and clear to send (CTS).

In the example of FIG. 7, in order to efficiently use a collision avoidance mechanism, short signaling packet such as RTS and CTS may be used. RST/CTS between two STAs may be enabled to be overheard by peripheral STAs such that the peripheral STAs confirm information transmission between the two STAs. For example, if a transmission STA transmits an RTS frame to a reception STA, the reception STA transmits a CTS frame to peripheral UEs to inform the peripheral UEs that the reception STA receives data.

FIG. 8(a) shows a method of solving a hidden node problem. Assume that both the STA A and the STA C attempt to transmit data to the STA B. If the STA A transmits the RTS to the STA B, the STA B transmits the CTS to the peripheral STA A and C. As a result, the STA C waits until data transmission of the STA A and the STA B is finished, thereby avoiding collision.

FIG. 8(b) shows a method of solving an exposed node problem. The STA C may overhear RTS/CTS transmission between the STA A and the STA B and determine that collision does not occur even when the STA C transmits data to another STA (e.g., the STA D). That is, the STA B transmits the RTS to all peripheral UEs and transmits the CTS only to the STA A having data to be actually transmitted. Since the STA C receives the RTS but does not receive the CTS of the STA A, it can be confirmed that the STA A is outside carrier sensing of the STA C.

2.4 Power Management

As described above, in a WLAN system, channel sensing should be performed before an STA performs transmission and reception. When the channel is always sensed, continuous power consumption of the STA is caused. Power consumption in a reception state is not substantially different from power consumption in a transmission state and continuously maintaining the reception state imposes a burden on an STA with limited power (that is, operated by a battery). Accordingly, if a reception standby state is maintained such that the STA continuously senses the channel, power is inefficiently consumed without any special advantage in terms of WLAN throughput. In order to solve such a problem, in a WLAN system, a power management (PM) mode of the STA is supported.

The PM mode of the STA is divided into an active mode and a power save (PS) mode. The STA fundamentally operates in an active mode. The STA which operates in the active mode is maintained in an awake state. The awake state refers to a state in which normal operation such as frame transmission and reception or channel scanning is possible. The STA which operates in the PS mode operates while switching between a sleep state or an awake state. The STA which operates in the sleep state operates with minimum power and does not perform frame transmission and reception or channel scanning.

Since power consumption is reduced as the sleep state of the STA is increased, the operation period of the STA is increased. However, since frame transmission and reception is impossible in the sleep state, the STA may not unconditionally operate in the sleep state. If a frame to be transmitted from the STA, which operates in the sleep state, to the AP is present, the STA may be switched to the awake state to transmit the frame. If a frame to be transmitted from the AP to the STA is present, the STA in the sleep state may not receive the frame and may not confirm that the frame to be received is present. Accordingly, the STA needs to perform an operation for switching to the awake state according to a specific period in order to confirm presence of the frame to be transmitted thereto (to receive the frame if the frame to be transmitted is present).

FIG. 9 is a diagram illustrating power management operation.

Referring to FIG. 9, an AP 210 transmits beacon frames to STAs within a BSS at a predetermined period (S211, S212, S213, S214, S215 and S216). The beacon frame includes a traffic indication map (TIM) information element. The TIM information element includes information indicating that buffered traffic for STAs associated with the AP 210 is present and the AP 210 will transmit a frame. The TIM element includes a TIM used to indicate a unicast frame or a delivery traffic indication map (DTIM) used to indicate a multicast or broadcast frame.

The AP 210 may transmit the DTIM once whenever the beacon frame is transmitted three times. An STA1 220 and an STA2 222 operate in the PS mode. The STA1 220 and the STA2 222 may be switched from the sleep state to the awake state at a predetermined wakeup interval to receive a TIM element transmitted by the AP 210. Each STA may compute a time to switch to the awake state based on a local clock thereof. In the example of FIG. 9, assume that the clock of the STA matches the clock of the AP.

For example, the predetermined awake interval may be set such that the STA1 220 is switched to the awake state every beacon interval to receive a TIM element. Accordingly, the STA1 220 may be switched to the awake state (S211) when the AP 210 first transmits the beacon frame (S211). The STA1 220 may receive the beacon frame and acquire the TIM element. If the acquired TIM element indicates that a frame to be transmitted to the STA1 220 is present, the STA1 220 may transmit, to the AP 210, a power save-Poll (PS-Poll) frame for requesting frame transmission from the AP 210 (S221 a). The AP 210 may transmit the frame to the STA1 220 in correspondence with the PS-Poll frame (S231). The STA1 220 which completes frame reception is switched to the sleep state.

When the AP 210 secondly transmits the beacon frame, since another device access the medium and thus the medium is busy, the AP 210 may not transmit the beacon frame at an accurate beacon interval and may transmit the beacon frame at a delayed time (S212). In this case, the operation mode of the STA1 220 is switched to the awake state according to the beacon interval but the delayed beacon frame is not received. Therefore, the operation mode of the STA1 220 is switched to the sleep state again (S222).

When the AP 210 thirdly transmits the beacon frame, the beacon frame may include a TIM element set to a DTIM. Since the medium is busy, the AP 210 transmits the beacon frame at a delayed time (S213). The STA1 220 is switched to the awake state according to the beacon interval and may acquire the DTIM via the beacon frame transmitted by the AP 210. Assume that the DTIM acquired by the STA1 220 indicates that a frame to be transmitted to the STA1 220 is not present and a frame for another STA is present. In this case, the STA1 220 may confirm that a frame transmitted thereby is not present and may be switched to the sleep state again. The AP 210 transmits the beacon frame and then transmits the frame to the STA (S232).

The AP 210 fourthly transmits the beacon frame (S214). Since the STA1 220 cannot acquire information indicating that buffered traffic therefor is present via reception of the TIM element twice, the wakeup interval for receiving the TIM element may be controlled. Alternatively, if signaling information for controlling the wakeup interval of the STA1 220 is included in the beacon frame transmitted by the AP 210, the wakeup interval value of the STA1 220 may be controlled. In the present example, the STA1 220 may change switching of the operation state for receiving the TIM element every beacon interval to switching of the operation state every three beacon intervals. Accordingly, since the STA1 220 is maintained in the sleep state when the AP 210 transmits the fourth beacon frame (S214) and transmits the fifth beacon frame (S215), the TIM element cannot be acquired.

When the AP 210 sixthly transmits the beacon frame (S216), the STA1 220 may be switched to the awake state to acquire the TIM element included in the beacon frame (S224). Since the TIM element is a DTIM indicating that a broadcast frame is present, the STA1 220 may not transmit the PS-Poll frame to the AP 210 but may receive a broadcast frame transmitted by the AP 210 (S234). The wakeup interval set in the STA2 230 may be set to be greater than that of the STA1 220. Accordingly, the STA2 230 may be switched to the awake state to receive the TIM element (S241), when the AP 210 fifthly transmits the beacon frame (S215). The STA2 230 may confirm that a frame to be transmitted thereto is present via the TIM element and transmits the PS-Poll frame to the AP 210 (S241 a) in order to request frame transmission. The AP 210 may transmit the frame to the STA2 230 in correspondence with the PS-Poll frame (S233).

For PM management shown in FIG. 9, a TIM element includes a TIM indicating whether a frame to be transmitted to an STA is present and a DTIM indicating whether a broadcast/multicast frame is present. The DTIM may be implemented by setting a field of the TIM element.

FIGS. 10 to 12 are diagrams illustrating operation of a station (STA) which receives a traffic indication map (TIM).

Referring to FIG. 10, an STA may be switched from a sleep state to an awake state in order to receive a beacon frame including a TIM from an AP and interpret the received TIM element to confirm that buffered traffic to be transmitted thereto is present. The STA may contend with other STAs for medium access for transmitting a PS-Poll frame and then transmit the PS-Poll frame in order to request data frame transmission from the AP. The AP which receives the PS-Poll frame transmitted by the STA may transmit the frame to the STA. The STA may receive the data frame and transmit an ACK frame to the AP. Thereafter, the STA may be switched to the sleep state again.

As shown in FIG. 10, the AP may receive the PS-Poll frame from the STA and then operate according to an immediate response method for transmitting a data frame after a predetermined time (e.g., a short inter-frame space (SIFS)). If the AP does not prepare a data frame to be transmitted to the STA during the SIFS after receiving the PS-Poll frame, the AP may operate according to a deferred response method, which will be described with reference to FIG. 11.

In the example of FIG. 11, operation for switching the STA from the sleep state to the awake state, receiving a TIM from the AP, contending and transmitting a PS-Poll frame to the AP is equal to that of FIG. 10. If the data frame is not prepared during the SIFS even when the AP receives the PS-Poll frame, the data frame is not transmitted but an ACK frame may be transmitted to the STA. If the data frame is prepared after transmitting the ACK frame, the AP may contend and transmit the data frame to the STA. The STA may transmit the ACK frame indicating that the data frame has been successfully received to the AP and may be switched to the sleep state.

FIG. 12 shows an example in which the AP transmits the DTIM. The STAs may be switched from the sleep state to the awake state in order to receive the beacon frame including the DTIM element from the AP. The STA may confirm that a multicast/broadcast frame will be transmitted via the received DTIM. The AP may immediately transmit data (that is, a multicast/broadcast frame) without PS-Poll frame transmission and reception after transmitting the beacon frame including the DTIM. The STAs may receive data in the awake state after receiving the beacon frame including the DTIM and may be switched to the sleep state again after completing data reception.

2.5 TIM Structure

In the PM mode management method based on the TIM (or DTIM) protocol described with reference to FIGS. 9 to 12, the STAs may confirm whether a data frame to be transmitted thereto is present via STA identification included in the TIM element. The STA identification may be related to an association identifier (AID) assigned to the STA upon association with the AP.

The AID is used as a unique identifier for each STA within one BSS. For example, in a current WLAN system, the AID may be one of values of 1 to 2007. In a currently defined WLAN system, 14 bits are assigned to the AID in a frame transmitted by the AP and/or the STA. Although up to 16383 may be assigned as the AID value, 2008 to 16383 may be reserved.

The TIM element according to an existing definition is not appropriately applied to an M2M application in which a large number (e.g., more than 2007) of STAs is associated with one AP. If the existing TIM structure extends without change, the size of the TIM bitmap is too large to be supported in an existing frame format and to be suitable for M2M communication considering an application with a low transfer rate. In addition, in M2M communication, it is predicted that the number of STAs, in which a reception data frame is present during one beacon period, is very small. Accordingly, in M2M communication, since the size of the TIM bitmap is increased but most bits have a value of 0, there is a need for technology for efficiently compressing the bitmap.

As an existing bitmap compression technology, a method of omitting 0 which continuously appears at a front part of a bitmap and defining an offset (or a start point) is provided. However, if the number of STAs in which a buffered frame is present is small but a difference between the AID values of the STAs is large, compression efficiency is bad. For example, if only frames to be transmitted to only two STAs respectively having AID values of 10 and 2000 are buffered, the length of the compressed bitmap is 1990 but all bits other than both ends have a value of 0. If the number of STAs which may be associated with one AP is small, bitmap compression inefficiency is not problematic but, if the number of STAs is increased, bitmap compression inefficiency deteriorates overall system performance.

As a method of solving this problem, AIDs may be divided into several groups to more efficiently perform data transmission. A specific group ID (GID) is assigned to each group. AIDs assigned based on the group will be described with reference to FIG. 13.

FIG. 13(a) shows an example of AIDs assigned based on a group. In the example of FIG. 13(a), several bits of a front part of the AID bitmap may be used to indicate the GID. For example, four DIDs may be expressed by the first two bits of the AID of the AID bitmap. If the total length of the AID bitmap is N bits, the first two bits (B1 and B2) indicate the GID of the AID.

FIG. 13(a) shows another example of AIDs assigned based on a group. In the example of FIG. 13(b), the GID may be assigned according to the location of the AID. At this time, the AIDs using the same GID may be expressed by an offset and a length value. For example, if GID 1 is expressed by an offset A and a length B, this means that AIDs of A to A+B−1 on the bitmap have GID 1. For example, in the example of FIG. 13(b), assume that all AIDs of 1 to N4 are divided into four groups. In this case, AIDs belonging to GID 1 are 1 to N1 and may be expressed by an offset 1 and a length N1. AIDs belonging to GID2 may be expressed by an offset N1+1 and a length N2−N1+1, AIDs belonging to GID 3 may be expressed by an offset N2+1 and a length N3−N2+1, and AIDs belonging to GID 4 may be expressed by an offset N3+1 and a length N4−N3+1.

If the AIDs assigned based on the group are introduced, channel access is allowed at a time interval which is changed according to the GID to solve lack of TIM elements for a large number of STAs and to efficiently perform data transmission and reception. For example, only channel access of STA(s) corresponding to a specific group may be granted during a specific time interval and channel access of the remaining STA(s) may be restricted. A predetermined time interval at which only access of specific STA(s) is granted may also be referred to as a restricted access window (RAW).

Channel access according to GID will be described with reference to FIG. 13(c). FIG. 13(c) shows a channel access mechanism according to a beacon interval if the AIDs are divided into three groups. At a first beacon interval (or a first RAW), channel access of STAs belonging to GID 1 is granted but channel access of STAs belonging to other GIDs is not granted. For such implementation, the first beacon includes a TIM element for AIDs corresponding to GID 1. A second beacon frame includes a TIM element for AIDs corresponding to GID 2 and thus only channel access of the STAs corresponding to the AIDs belonging to GID 2 is granted during the second beacon interval (or the second RAW). A third beacon frame includes a TIM element for AIDs corresponding to GID 3 and thus only channel access of the STAs corresponding to the AIDs belonging to GID 3 is granted during the third beacon interval (or the third RAW). A fourth beacon frame includes a TIM element for AIDs corresponding to GID 1 and thus only channel access of the STAs corresponding to the AIDs belonging to GID 1 is granted during the fourth beacon interval (or the fourth RAW). Only channel access of the STAs corresponding to a specific group indicated by the TIM included in the beacon frame may be granted even in fifth and subsequent beacon intervals (or fifth and subsequent RAWs).

Although the order of GIDs allowed according to the beacon interval is cyclic or periodic in FIG. 13(c), the present invention is not limited thereto. That is, by including only AID(s) belonging to specific GID(s) in the TIM elements, only channel access of STA(s) corresponding to the specific AID(s) may be granted during a specific time interval (e.g., a specific RAW) and channel access of the remaining STA(s) may not be granted.

The above-described group based AID assignment method may also be referred to as a hierarchical structure of a TIM. That is, an entire AID space may be divided into a plurality of blocks and only channel access of STA(s) corresponding to a specific block having a non-zero value (that is, STAs of a specific group) may be granted. A TIM having a large size is divided into small blocks/groups such that the STA easily maintains TIM information and easily manages blocks/groups according to class, QoS or usage of the STA. Although a 2-level layer is shown in the example of FIG. 13, a TIM of a hierarchical structure having two or more levels may be constructed. For example, the entire AID space may be divided into a plurality of page groups, each page group may be divided into a plurality of blocks, and each block may be divided into a plurality of sub-blocks. In this case, as an extension of the example of FIG. 13(a), the first N1 bits of the AID bitmap indicate a paging ID (that is, a PID), the next N2 bits indicate a block ID, the next N3 bits indicate a sub-block ID, and the remaining bits indicate the STA bit location in the sub-block.

In the following examples of the present invention, various methods of dividing and managing STAs (or AIDs assigned to the STAs) on a predetermined hierarchical group basis are applied and the group based AID assignment method is not limited to the above examples.

2.6 Improved Channel Access Method

If AIDs are assigned/managed based on a group, STAs belonging to a specific group may use a channel only at a “group channel access interval (or RAW)” assigned to the group. If an STA supports an M2M application, traffic for the STA may have a property which may be generated at a long period (e.g., several tens of minutes or several hours). Since such an STA does not need to be in the awake state frequently, the STA may be in the sleep mode for g a long period of time and be occasionally switched to the awake state (that is, the awake interval of the STA may be set to be long). An STA having a long wakeup interval may be referred to as an STA which operates in a “long-sleeper” or “long-sleep” mode. The case in which the wakeup interval is set to be long is not limited to M2M communication and the wakeup interval may be set to be long according to the state of the STA or surroundings of the STA even in normal WLAN operation.

If the wakeup interval is set, the STA may determine whether a local clock thereof exceeds the wakeup interval. However, since the local clock of the STA generally uses a cheap oscillator, an error probability is high. In addition, if the STA operates in long-sleep mode, the error may be increased with time. Accordingly, time synchronization of the STA which occasionally wakes up may not match time synchronization of the AP. For example, although the STA computes when the STA may receive the beacon frame to be switched to the awake state, the STA may not actually receive the beacon frame from the AP at that timing. That is, due to clock drift, the STA may miss the beacon frame and such a problem may frequently occur if the STA operates in the long sleep mode.

FIGS. 14 to 16 are diagrams showing examples of operation of an STA if a group channel access interval is set.

In the example of FIG. 14, STA3 may belong to group 3 (that is, GID=3), wake up at a channel access interval assigned to group 1 and perform PS-Poll for requesting frame transmission from the AP. The AP which receives PS-Poll from the STA transmits an ACK frame to STA3. If buffered data to be transmitted to STA3 is present, the AP may provide information indicating that data to be transmitted is present via the ACK frame. For example, the value of a “More Data” field (or an MD field) having a size of 1 bit included in the ACK frame may be set to 1 (that is, MD=1) to indicate the above information.

Since a time when STA3 transmits PS-Poll belongs to the channel access interval for group 1, even if data to be transmitted to STA3 is present, the AP does not immediately transmit data after transmitting the ACK frame but transmits data to STA3 at a channel access interval (GID 3 channel access of FIG. 14) assigned to group 3 to which STA3 belongs.

Since STA3 receives the ACK frame set to MD=1 from the AP, STA3 continuously waits for transmission of data from the AP. That is, in the example of FIG. 14, since STA3 cannot receive the beacon frame immediately after waking up, STA3 transmits PS-Poll to the AP on the assumption that a time when STA3 wakes up corresponds to the channel access interval assigned to the group, to which STA3 belongs, according to computation based on the local clock thereof and data to be transmitted thereto is present. Alternatively, since STA3 operates in the long-sleep mode, on the assumption that time synchronization is not performed, if the data to be transmitted thereto is present, STA3 may transmit PS-Poll to the AP in order to receive the data. Since the ACK frame received by STA3 from the AP indicates that data to be transmitted to STA3 is present, STA3 continuously waits for data reception under the assumption of the interval in which channel access thereof is granted. STA3 unnecessarily consumes power even when data reception is not allowed, until time synchronization is appropriately performed from information included in a next beacon frame.

In particular, if STA3 operates in the long-sleep mode, the beacon frame may frequently not be received, CCA may be performed even at the channel access interval, to which STA2 does not belong, thereby causing unnecessary power consumption.

Next, in the example of FIG. 15, the beacon frame is missed when the STA having GID 1 (that is, belonging to group 1) wakes up. That is, the STA which does not receive the beacon frame including the GID (or PID) assigned thereto is continuously in the awake state until the beacon frame including the GID (or PID) thereof is received. That is, although the STA wakes up at channel access interval assigned thereto, the STA cannot confirm whether the GID (or PID) thereof is included in the TIM transmitted via the beacon frame and thus cannot confirm whether the timing corresponds to the channel access interval assigned to the group thereof.

In the example of FIG. 15, the STA which is switched from the sleep state to the awake state is continuously in the awake state until the fourth beacon frame including the GID (that is, GID 1) thereof is received after the first beacon frame has been missed, thereby causing unnecessary power consumption. As a result, after unnecessary power consumption, the STA may receive the beacon frame including GID 1 and then may perform RTS transmission, CTS reception, data frame transmission and ACK reception.

FIG. 16 shows the case in which an STA wakes up at a channel access interval for another group. For example, the STA having GID 3 may wake up at the channel access interval for GID 1. That is, the STA having GID 3 unnecessarily consumes power until the beacon frame having the GID thereof is received after waking up. If a TIM indicating GID 3 is received via a third beacon frame, the STA may recognize the channel access interval for the group thereof and perform data transmission and ACK reception after CCA through RTS and CTS.

3. First Proposed Method for CFO Estimation

As interest in future Wi-Fi and demand for improvement of throughput and QoE (quality of experience) after 802.11ac increase, it is necessary to define a new frame format for future WLAN systems. The most important part in a new frame format is a preamble part because design of a preamble used for synchronization, channel tracking, channel estimation, adaptive gain control (AGC) and the like may directly affect system performance.

In the future Wi-Fi system in which a large number of APs and STAs simultaneously access and attempt data transmission and reception, system performance may be limited when legacy preamble design is employed. That is, if each preamble block (e.g., a short training field (STF) in charge of AGC, CFO estimation/compensation, timing control and the like or a long training field (LTF) in charge of channel estimation/compensation, residual CFO compensation and the like) executes only the function thereof defined in the legacy preamble structure, frame length increases, causing overhead. Accordingly, if a specific preamble block can support various functions in addition to the function designated therefor, an efficient frame structure can be designed.

Furthermore, since the future Wi-Fi system considers data transmission in outdoor environments as well as indoor environments, the preamble structure may need to be designed differently depending on environments. Although design of a unified preamble format independent of environment variation can aid in system implementation and operation, of course, it is desirable that preamble design be adapted to system environment.

Preamble design for efficiently supporting various functions is described hereinafter. For convenience, a new WLAN system is referred to as an HE (High Efficiency) system and a frame and a PPDU (PLCP (Physical Layer Convergence Procedure) Protocol Data Unit) of the HE system are respectively referred to as an HE frame and an HE PPDU. However, it is obvious to those skilled in the art that the proposed preamble is applicable to other WLAN systems and cellular systems in addition to the HE system.

The following table 1 shows OFDM numerology which is a premise of a pilot sequence transmission method described below. Table 1 shows an example of new OFDM numerology proposed in the HE system and numerals and items shown in Table 1 are merely examples and other values may be applied. Table 1 is based on the assumption that FFT having a size four times the legacy one is applied to a given BW and 3 DCs are used per BW.

TABLE 1 Parameter CBW20 CBW40 CBW80 CBW80 + 80 CBW160 Description N_(FFT) 256 512 1024 1024 2048 FFT size N_(SD) 238 492 1002 1002 2004 Number of complex data numbers per frequency segment N_(SP) 4 6 8 8 16 Number of pilot values per frequency segment N_(ST) 242 498 1010 1010 2020 Total number of subcarriers per frequency segment. See NOTE. N_(SR) 122 250 506 506 1018 Highest data subcarrier index per frequency segment N_(Seg) 1 1 1 2 1 Number of frequency segments Δ_(F) 312.5 kHz Subcarrier frequency Spacing for non-HE portion Δ_(F) _(—) _(HE) 78.125 kHz Subcarrier frequency Spacing for HE portion T_(DFT) 3.2 μs IDFT/DFT period for non-HE portion T_(DFT) _(—) _(HE) 12.8 μs IDFT/DFT period for HE portion T_(GI) 0.8 μs = T_(DFT)/4 Guard interval duration for non- HE portion T_(GI) _(—) _(HE) 3.2 μs = T_(DFT) _(—) _(HE)/4 Guard interval duration for HE portion T_(GI2) 1.6 μs Double guard interval for non-HE portion T_(GIS) _(—) _(HE) 0.8 μs = T_(DFT) _(—) _(HE)/16 [Alternative: 0.4 μs ( 1/32 CP)] Short guard interval Duration (used only for HE data) T_(SYML) 4 μs = T_(DFT) + T_(GI) Long GI symbol interval for non-HE portion T_(SYML) _(—) _(HE) 16 μs = T_(DFT) _(—) _(HE) + T_(GI) _(—) _(HE) Long GI symbol interval for HE portion T_(SYMS) _(—) _(HE) 13.6 μs = T_(DFT) _(—) _(HE) + T_(GIS) _(—) _(HE) [Alternative: 13.2 μs Short GI symbol (with 1/32 CP)] interval (used only for HE data) T_(SYM) T_(SYML) or T_(SYMS) depending on the GI used Symbol interval for non-HE portion T_(SYM) _(—) _(HE) T_(SYML) _(—) _(HE) or T_(SYMS) _(—) _(HE) depending on the GI used Symbol interval for HE portion T_(L-STF) 8 μs = 10 * T_(DFT)/4 Non-HE Short Training field duration T_(L-LTF) 8 μs = 2 × T_(DFT) + T_(GI2) Non-HE Long Training field duration T_(L-SIG) 4 μs = T_(SYML) Non-HE SIGNAL field duration T_(HE-SIGA) 12.8 μs = 2(T_(SYML) + 3T_(GI)) in HE-PPDU format-1 or HE Signal A field T_(SYML) _(—) _(HE) in HE-PPDU format-2 duration and HE-PPDU format-3 T_(HE-STF) T_(SYML) _(—) _(HE) HE Short Training field duration T_(HE-LTF) T_(SYML) _(—) _(HE) Duration of each HE LTF symbol T_(HE-SIGB) T_(SYML) _(—) _(HE) HE Signal B field duration N_(service) 16 Number of bits in the SERVICE field N_(tail)  6 Number of tail bits per BCC encoder NOTE N_(ST) = N_(SD) + N_(SP)

FIG. 17 is a diagram illustrating frame structures related to an embodiment of the present invention. As illustrated in FIGS. 17(a), 17(b) and 17(c), various frame structures can be configured, and a proposed pilot sequence transmission method is related to an HE-STF (High Efficiency Short Training Field) in a preamble in a frame structure.

FIGS. 18 and 19 are diagrams illustrating frame structures according to another embodiment of the present invention and constellations thereof. Specifically, FIG. 18 (a) illustrates a time-domain frame structure of a high throughput (HT) system based on 802.11n. In FIG. 18 (a), L-SIG and HT-SIG indicates a legacy signal field and a high throughput signal field, respectively. Assuming that one OFDM symbol length is 4 us, the L-SIG corresponds to a single OFDM symbol and the HT-SIG corresponds to two OFDM symbols.

Meanwhile, assuming that data transmission is performed based on the frame structure shown in FIG. 18 (a), system information can be mapped to constellations shown in FIG. 18 and then transmitted to a UE.

FIG. 19 (a) shows a frame structure of a very high throughput (VHT) system based on 802.11ac. Similar to FIG. 18, in the VHT system, system information is mapped to constellations shown in FIG. 19 (b) and then transmitted to a UE using L-SIG and VHT-SIG-A fields shown in FIG. 19 (a).

FIG. 20 is a diagram illustrating frequency-domain pilot signals related to the proposed embodiments.

After receiving such fields as L-SIG, HT-SIG, and VHT-SIG-A from a BS (or transmission module) as described with reference to FIGS. 18 and 19, a UE (or reception module) performs Fast Fourier Transform (FFT) operation. The results of the operation can be expressed as shown in FIG. 20.

FIG. 20 shows converted frequency-domain pilot signals in each OFDM symbol. In FIG. 20, a signal received through each subcarrier can be expressed as shown in Equation 1. r _(k) ^(n) =H _(k) ^(n) s _(k) ^(n)  [Equation 1]

In Equation 1, k denotes a subcarrier index and n denotes an OFDM symbol index. In addition, H_(k) ^(n) indicates a channel between an n^(th) OFDM symbol and a k^(th) subcarrier. Assuming that a data signal transmitted through H_(k) ^(n) is s_(k) ^(n), a received signal can be expressed as r_(k) in Equation 1.

Meanwhile, as shown in FIG. 20, some subcarriers include guard intervals or direct current (DC) components and such carriers are set to null without loading data signals. On the other hand, in case of subcarriers in which data signals are loaded, a set of their indices is defined as C.

Before describing the proposed CFO estimation method, the concept of a CFO is explained hereinbelow. The CFO (carrier frequency offset) occurs due to performance of oscillators included in both a transmission module and a reception module or Doppler effects. The CFO can be divided into an integer part and a fractional part (for example, if the CFO has a value of 2.5, the integer part is 2 and the fractional part is 0.5). A subcarrier is circular shifted by the integer part of the CFO, but the fractional part of the CFO causes interference between subcarriers.

In the HT and VHT systems, a reception module estimates a CFO value using an L-STF field and an L-LTF field. After the CFO estimation, the estimated result is applied to a received OFDM symbol. By doing so, the effect of the CFO is eliminated as shown in Equation 2.

$\begin{matrix} \begin{matrix} {{{D\left( {- \hat{\epsilon}} \right)}y} = {{D\left( {- \hat{\epsilon}} \right)}\left( {{{D(\epsilon)}x} + n} \right)}} \\ {= {{{D\left( {\epsilon - \hat{\epsilon}} \right)}x} + n^{\prime}}} \\ {= {{{D\left( {\Delta\;\epsilon} \right)}x} + n^{\prime}}} \end{matrix} & \left\lbrack {{Equation}\mspace{14mu} 2} \right\rbrack \end{matrix}$

In Equation 2, ϵ indicates an actual CFO value and {circumflex over (ϵ)} indicates an estimated CFO value. In addition, y indicates a received signal vector when the CFO is present, x indicates a received signal vector when the CFO is not present, and n indicates a noise vector. A diagonal matrix D of Equation 2 is defined as shown in Equation 3.

$\begin{matrix} {{D(\epsilon)} = \begin{bmatrix} 1 & 0 & \; & 0 \\ 0 & e^{j\; 2\;\pi\;{\epsilon/N}} & \ddots & \; \\ \; & \ddots & \ddots & 0 \\ 0 & \; & 0 & e^{j\; 2\;\pi\;{{\epsilon{({N - 1})}}/N}} \end{bmatrix}} & \left\lbrack {{Equation}\mspace{14mu} 3} \right\rbrack \end{matrix}$

If the reception module perfectly estimates the CFO value using the L-SFT and the L-LTF (ϵ={circumflex over (ϵ)}), the reception module can perfectly eliminate the CFO from the received signal using Equation 2 and Equation 3 (Δϵ=0). However, since the CFO is slightly changed depending on time, it is difficult to perfectly estimate the CFO value (ϵ≠{circumflex over (ϵ)}). Thus, a residual CFO can be defined as shown in Equation 4. Δϵ=ϵ−{circumflex over (ϵ)}+{tilde over (ϵ)}  [Equation 4]

In Equation 4, {tilde over (ϵ)} indicates a CFO value changed depending on time. To re-estimate the residual CFO, the reception module utilizes pilot signals included in the L-SIG and HT-SIG. In the HT system, the residual CFO is estimated using four pilot signals. However, since the HT system has a relatively small number of pilot signals, performance of the CFO estimation is significantly decreased in case of a low SNR. That is, the number of pilot signal needs to be increased to overcome such a problem but it may cause throughput reduction as a trade-off. Therefore, a CFO estimation method for minimizing performance degradation in case of a low SNR while maintaining an HT system structure needs to be developed.

Hereinafter, a CFO method according to the present invention will be described with reference to FIGS. 21 to 23. According to proposed embodiments, the reception module can estimate the CFO in a blind manner, i.e., using a data signal instead of a pilot signal.

FIGS. 21 and 22 are diagrams for explaining a proposed CFO estimation method. In FIGS. 21 and 22, it is assumed that data transmission is performed according to a binary phase shift keying (BPSK) scheme or a quadrature phase shift keying (QBPSK) scheme.

According to the proposed CFO estimation method, y_(k) ^(n), which reflects a received signal in two consecutive OFDM symbols based on Equation 1, can be defined as shown in Equation 5. y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*,k∈C,n=1,2, . . . ,L  [Equation 5]

In Equation 5, L is defined as (the number of total OFDM symbols to which the proposed CFO estimation method is applied—1). For example, when two OFDM symbols are used as shown in FIG. 20, L is set to 1. When three OFDM symbols are used for the L-SIG and the HT-SIG shown in FIG. 18 (a), L is set to 3.

Hereinafter, Equation 5 is described in detail. In the case of two consecutive OFDM symbols (i.e., n^(th) OFDM symbol and (n+1)^(th) OFDM symbol), a channel is not rapidly changed. In other words, in Equation 5, y_(k) ^(n) is defined on the assumption that the two consecutive OFDM symbols have the same channel.

According to the proposed CFO estimation method, a process shown in Equation 6 below is performed after calculation of y_(k) ^(n).

$\begin{matrix} {z_{k}^{n} = \left\{ \begin{matrix} y_{k}^{n} & {{{if}\mspace{14mu}{real}\mspace{11mu}\left( y_{k}^{n} \right)} \geq 0} \\ {- y_{k}^{n}} & {{{if}\mspace{14mu}{real}\mspace{11mu}\left( y_{k}^{n} \right)} < 0} \end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 6} \right\rbrack \end{matrix}$

In Equation 6, z_(k) ^(n) is determined by a sign of a real part of y_(k) ^(n). In addition, Equation 7 below shows a process for determining a final residual CFO.

$\begin{matrix} {{\Delta\;\hat{\epsilon}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in C}z_{k}^{n}}} \right)} \times \frac{N}{2\;{\pi\left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 7} \right\rbrack \end{matrix}$

In equation 7, Δ{circumflex over (ϵ)} indicates a finally calculated residual CFO value, and N and N_(g) indicate an OFDM symbol length and a cyclic prefix (CP) length, respectively. It can be seen from Equation 7 that the processes described in Equation 5 and Equation 6 are performed with respect to the entirety of the set C consisting of subcarriers where data are loaded.

Hereinafter, processes for Equations 5 to 7 are described in detail. Assuming that n is equal to 1 in Equation 5, y_(k) ¹ can be expressed as shown in Equation 8.

$\begin{matrix} \begin{matrix} {y_{k}^{1}\overset{\Delta}{=}{r_{k}^{2}\left( r_{k}^{1} \right)}^{*}} \\ {\approx {H_{k}^{2}{s_{k}^{2}\left( {H_{k}^{1}s_{k}^{1}} \right)}^{*}e^{j\; 2\;\pi\;\Delta\;{{\epsilon{({N + N_{g}})}}/N}}}} \\ {= {{H_{k}^{2}\left( H_{k}^{1} \right)}^{*}{s_{k}^{2}\left( s_{k}^{1} \right)}^{*}e^{j\; 2\;\pi\;\Delta\;{{\epsilon{({N + N_{g}})}}/N}}}} \\ {\approx {\rho{H_{k}^{1}}^{2}{{sign}\left( {s_{k}^{2}\left( s_{k}^{1} \right)}^{*} \right)}e^{j\; 2\;\pi\;\Delta\;{{\epsilon{({N + N_{g}})}}/N}}}} \end{matrix} & \left\lbrack {{Equation}\mspace{14mu} 8} \right\rbrack \end{matrix}$

In Equation 8, ρ indicates a power component of the data signal s_(k) ^(n). In addition, a function sign(a) has a value of 1 when a variable a has a positive sign and a value of −1 when the variable a has a negative sign. Meanwhile, approximation in the second line of Equation 8 is achieved based on the assumption that interference between subcarriers caused by the residual CFO can be ignored. Moreover, approximation in the fourth line is achieved on the assumption that channels H_(k) ¹ and H_(k) ² in the two OFDM symbols are equal to each other. Considering that the BPSK scheme is applied together with the above assumptions, an equation of s_(k) ²(s_(k) ¹)*=ρsign(s_(k) ²(s_(k) ¹)*) is always satisfied.

In Equation 8, when an equation of ρsign(s_(k) ²(s_(k) ¹)*)=ρ is satisfied, y_(k) ¹ is defined as shown in Equation 9. y _(k) ¹ ≈ρ|H _(k) ¹|² e ^(j2πΔϵ(N+N) ^(g) ^()/N)  [Equation 9]

Referring to FIG. 21, it can be seen that a component ρ|H_(k) ¹|² corresponds to a radius of the illustrated circle and a component 2πΔϵ(N+N_(g))/N corresponds to a phase value of the illustrated point. In this case, a phase value of y_(k) ¹ is a function of the residual CFO (Δ{circumflex over (ϵ)}) and the value is proportional to the residual CFO value. For example, if the residual CFO value is 0, the phase of y_(k) ¹ is also 0. If the phase of y_(k) ¹ is smaller than 2π, a ratio of the residual CFO to the phase of y_(k) ¹ is 1:1. Thus, it is possible to estimate the residual CFO value from the phase of y_(k) ¹.

On the other hand, when an equation of is ρsign(s_(k) ²(s_(k) ¹)*)=−ρ is satisfied, Equation 9 is expressed as shown in Equation 22 instead of Equation 21. If the reception module is aware of ρsign(s_(k) ²(s_(k) ¹)*)=−ρ, the reception module can estimate the residual CFO from the phase of y_(k) ¹ through a simple calculation. However, such a calculation is disadvantageous in that the reception module should know received data before performing the calculation. Moreover, according to the proposed CFO method, since the CFO estimation is performed in a blind manner without the use of a pilot signal, whether a value of sign(s_(k) ²(s_(k) ¹)*) is positive or negative should be first determined to accurately estimate the residual CFO.

To solve the above-mentioned problem, a case where the residual CFO value is relatively small compared to the total CFO is considered. In other words, first of all, the reception module may estimate the CFO value using a preamble part such as the L-STF and the L-LTF and then estimate the residual CFO value based on the L-SIG and the HT-SIG. When the CFO value is approximately estimated through the primary CFO estimation process as described above, a phase of the residual CFO has a relatively small value and thus, a range of the phase of y_(k) ¹ also does not have a large value. Accordingly, Equation 10 can be deduced as follows. sign(s _(k) ²(s _(k) ¹)*)=1 if angle(y _(k) ¹)ϵ{first quadrant,fourth quadrant} sign(s _(k) ²(s _(k) ¹)*)=−1 if angle(y _(k) ¹)ϵ{second quadrant,third quadrant}  [Equation 10]

If the phase of y_(k) ¹ belongs to the first quadrant or the fourth quadrant, the equation of ρsign(s_(k) ²(s_(k) ¹)*)=ρ is satisfied. On the contrary, if the phase of y_(k) ¹ belongs to the second quadrant or the third quadrant, the equation of ρsign(s_(k) ²(s_(k) ¹)*)=−ρ is satisfied. That is, Equation 6 can be explained by Equation 10, i.e., a residual CFO relationship. In other words, according to Equation 6, in the case of ρsign(s_(k) ²(s_(k) ¹)*)=−ρ, the phase of y_(k) ¹ is changed by π.

Meanwhile, when there is no noise, all phases of z_(k) ¹, k∈C, which are calculated according to Equation 6, have values of 2πΔϵ(N+N_(g))/N. That is, calculation results of all subcarriers are in a state of in-phase (or co-phase). According to Equation 7, the residual CFO can be accurately estimated when there is noise. In Equation 7, the phase and power of the signal are added all together, a robust result with respect to a dominant noise can be obtained.

According to the above-described method, the total CFO can be accurately measured by estimating the residual CFO in a blind manner. In addition, even when the SNR/SINR is low, overhead due to transmission of an additional pilot signal does not occur and thus, communication can be efficiently performed.

Meanwhile, according to an embodiment of the present invention, the CFO method can be applied to a subset C′ of the subcarrier set C where data is loaded. That is, although it can be seen in Equation 7 that the process for estimating the residual CFO is performed by adding phases of all samples, the residual CFO can be estimated using only some subcarriers instead of the total subcarrier samples. Accordingly, a subset of the set C where data signals are loaded is defined as the subset C′ and then the set C can be substituted with the subset C′ in Equation 7.

When a specific subcarrier is significantly faded, a size of data received through the corresponding subcarrier is also significantly decreased. In this case, compared to other data samples, the corresponding data sample is rarely attributed to the residual CFO estimation. In other words, the residual CFO can be estimated using only relatively large sizes of data samples. In addition, even if small sizes of samples are excluded, it does not cause performance degradation.

According to an embodiment generated by modifying the above-described embodiment, if the reception module knows sizes of data signals received through respective subcarriers, the reception module can arrange the subcarriers in descending order of sizes and then define subcarriers with sizes greater than a threshold as the subset C′. According to this embodiment, since the reception module may skip the processes described in Equations 5 to 7 (i.e., complexity associated with the processes described in Equations 5 to 7 may be reduced), performance degradation in the residual CFO estimation can also be minimized.

The present invention has been described with reference to cases where either the BPSK scheme or the QBPSK scheme is applied. However, the invention can be applied to a case where the BPSK scheme and the QBPSK scheme are alternately applied to each OFDM symbol. However, when two consecutive OFDM symbols are mapped to different constellations, the proposed method cannot be applied as it is because a product of two signals is not 1 or −1.

In this case, the residual CFO can be estimated according to Equation 11.

$\begin{matrix} {{y_{k}^{n}\overset{\Delta}{=}{{\overset{\sim}{r}}_{k}^{n + 1}\left( {\overset{\sim}{r}}_{k}^{n} \right)}^{*}},{k \in C},{n = 1},2,\ldots\mspace{14mu},L} & \left\lbrack {{Equation}\mspace{14mu} 11} \right\rbrack \\ {{\overset{\sim}{r}}_{k} = \left\{ \begin{matrix} {r_{k}e^{{- j}\;{\pi/2}}} & {{if}\mspace{14mu} s_{k}\mspace{14mu}{is}\mspace{14mu}{QBPSK}} \\ r_{k} & {otherwise} \end{matrix} \right.} & \; \end{matrix}$

Equation 5 can be substituted with Equation 11. In Equation 11, a phase of a signal received in the OFDM symbol where the QBPSK scheme is used is shifted by e^(−jπ/2) but a phase of a signal received in the OFDM symbol where the BPSK scheme is used is not shifted. According to Equation 11, when the BPSK and QBPSK are alternately used in a series of OFDM symbols, it is possible to obtain the same results as those in Equations 6 and 7.

FIG. 23 is a flowchart for explaining the proposed CFO estimation method. Specifically, FIG. 23 shows a time series flow of the CFO estimation method according to the aforementioned embodiments. Thus, it is apparent that although the aforementioned features described with reference to FIGS. 18 to 22 are not explicitly shown and described in FIG. 23, the features can be applied to the flowchart in FIG. 23 in the same or similar manner.

First, a transmission module transmits data to a reception module 52310. In this case, data is transmitted in a unit of frame, which is defined by an OFDM symbol and a subcarrier. In addition, the data is mapped to a specific constellation and then transmitted to the reception module. As such as a constellation, either BPSK or QBPSK may be used. Alternatively, the BPSK and QBPSK may be alternately used in a series of consecutive OFDM symbols.

Meanwhile, the reception module primarily estimate a CFO value based on a received signal [S2320]. Such a process is performed using a preamble part such as an L-STF, an L-LTF, and the like. However, since a CFO is changed depending on time, the CFO value estimated in the step S2320 may be inaccurate.

Therefore, the reception module estimate a residual CFO value to compensate the primarily estimated CFO value [S2330]. As described above, the reception module estimates the residual CFO value on the assumption that channels of received signals in two consecutive OFDM symbols are equal to each other. Specifically, the reception module calculate a product of the two received signal and then checks a sign of a real part of the product based on the assumption that the residual CFO value is smaller than the primarily estimated CFO value. If total subcarriers are in a state of in-phase, the reception module can obtain the residual CFO from a phase value calculated by adding results of all subcarriers.

Finally, the reception module can accurately decode the data transmitted from the transmission module by eliminating effects of the CFO estimated in the step S2320 and the residual CFO estimated in the step S2330 from the received signal.

4. Second Proposed Method for CFO Estimation

FIGS. 24 to 26 are diagrams illustrating a CFO estimation method according to a proposed embodiment. That is, the CFO estimation method when the BPSK and/or QBPSK scheme is used has been described above. Next, a description will be given of a CFO estimation method when data is transmitted using a quadrature phase shift keying (QPSK) scheme.

Similar to the process described in Equation 5, y_(k) ^(n), which reflects signals received in two consecutive OFDM symbols based on Equation 1, can be defined as shown in Equation 12. y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*,k∈C,n=1,2, . . . ,L  [Equation 12]

In Equation 12, L is defined as (the number of total OFDM symbols to which the proposed CFO estimation method is applied −1). For example, when two OFDM symbols are used as shown in FIG. 20, L is set to 1 (L=1). When three OFDM symbols are used similar to the L-SIG and the HT-SIG of FIG. 18 (a), L is set to 3 (L=3).

Hereinafter, Equation 12 is described in detail. In the case of two consecutive OFDM symbols (i.e., n^(th) OFDM symbol and (n+1)^(th) OFDM symbol), a channel is not rapidly changed. In other words, in Equation 12, y_(k) ^(n) is defined on the assumption that the two consecutive OFDM symbols have the same channel.

Thereafter, if y_(k) ^(n) is calculated at a reception module, a calculation can be performed as shown in Equation 12 below.

$\begin{matrix} {z_{k}^{n} = \left\{ \begin{matrix} y_{k}^{n} & {{{{if}\mspace{14mu}{{real}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\ {- y_{k}^{n}} & {{{{if}\mspace{14mu}{real}\;\left( y_{k}^{n} \right)} < 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\ {{- j} \times y_{k}^{n}} & {{{{if}\mspace{14mu}{{imag}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}} \\ {j \times y_{k}^{n}} & {{{{if}\mspace{14mu}{{imag}\left( y_{k}^{n} \right)}} < 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}} \end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 13} \right\rbrack \end{matrix}$

In equation 13, z_(k) ^(n) is determined by signs and magnitudes of a real part and an imaginary part of y_(k) ^(n). Although details will be described later, the four different cases in Equation 13 may respectively correspond to four different quadrants of the constellation. Thus, the function z_(k) ^(n) can be generated in different ways as shown in Equation 13 depending on a quadrant to which y_(k) ^(n) belongs.

Next, Equation 14 shows a process for determining a final residual CFO based on Equation 13.

$\begin{matrix} {{\Delta\;\hat{\epsilon}} = {{angle}\;\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in C}z_{k}^{n}}} \right) \times \frac{N}{2\;{\pi\left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 14} \right\rbrack \end{matrix}$

In Equation 14, Δ{circumflex over (ϵ)} indicates a final residual CFO value calculated at a reception module, and N and N_(g) indicate an OFDM symbol length and a cyclic prefix (CP) length, respectively. It can be seen from Equation 14 that the processes described in Equation 12 and Equation 13 are performed with respect to the entirety of the set C consisting of subcarriers where data are loaded.

Hereinafter, details of the processes in Equations 12 to 14 are described. First, y_(k) ¹ can be expressed as shown in Equation 15 according to the approximation procedure mentioned in Equation 8 y _(k) ¹ ≈|H _(k) ¹|² s _(k) ²(s _(k) ¹)*e ^(j2πΔϵ(N+N) ^(g) ^()/N)  [Equation 15]

When data transmission is performed using the QPSK scheme, s_(k) ²(s_(k) ¹)* can be distributed to one of the four points shown in FIG. 24. In other words, a phase of the product of data transmitted in the two consecutive OFDM symbols may be one of the four points of the constellation shown in FIG. 24. Meanwhile, in FIG. 24, the value of s_(k) ²(s_(k) ¹)* is defined as g_(k) ¹ according to Equation 16. g _(k) ¹ =s _(k) ²(s _(k) ¹)*[Equation 16]

It can be seen from FIG. 24 that a phase of g_(k) ¹ is one of {0, π/2, π, π/3}.

Meanwhile, if a residual CFO value to be measured at the reception module is relatively smaller than a total CFO value, y_(k) ¹ may have values shown in FIG. 25. That is, the residual CFO has a relatively smaller value than a CFO value firstly estimated based on the preamble part. Thus, it can be known that the phase of the function y_(k) ¹ used for residual CFO estimation does not have a relatively large value. Further, the phase of y_(k) ¹ may have values that are not significantly different from phase values at the four points of the QPSK constellation.

If the assumption that the residual CFO has a relatively small value is not established, a phase ambiguity problem may occur while the residual CFO is measured. However, based on the aforementioned assumption, it can be seen that the phase change of y_(k) ¹ due to the residual CFO is within the range of {0, π/2} (i.e., the first embodiment of FIG. 25), {π/2, π} (i.e., the second embodiment of FIG. 25), {π, 3π/2} (the third embodiment of FIG. 25), or {3π/2, 2π} (the fourth embodiment of FIG. 25).

Equation 17 can be deduced based on the above results.

$\begin{matrix} \left. y_{k}^{1}\Rightarrow\left\{ \begin{matrix} {◯1} & {{{{if}\mspace{14mu}{{real}\left( y_{k}^{1} \right)}} \geq 0},{{{real}\left( y_{k}^{1} \right)} \geq {{imag}\left( y_{k}^{1} \right)}}} \\ {◯3} & {{{{if}\mspace{14mu}{real}\;\left( y_{k}^{1} \right)} < 0},{{{real}\left( y_{k}^{1} \right)} \geq {{imag}\left( y_{k}^{1} \right)}}} \\ {◯2} & {{{{if}\mspace{14mu}{{imag}\left( y_{k}^{1} \right)}} \geq 0},{{{real}\left( y_{k}^{1} \right)} < {{imag}\left( y_{k}^{1} \right)}}} \\ {◯4} & {{{{if}\mspace{14mu}{{imag}\left( y_{k}^{1} \right)}} < 0},{{{real}\left( y_{k}^{1} \right)} < {{imag}\left( y_{k}^{1} \right)}}} \end{matrix} \right. \right. & \left\lbrack {{Equation}\mspace{14mu} 17} \right\rbrack \end{matrix}$

That is, it is possible to obtain a quadrant to which y_(k) ¹ belongs among the four quadrants of the constellation by comparing/analyzing a real part and an imaginary part of y_(k) ¹ according to Equation 17. For example, when y_(k) ¹ satisfies the condition of real(y_(k) ¹)≥0, real(y_(k) ¹)≥imag(y_(k) ¹), y_(k) ¹ corresponds to the first embodiment of FIGS. 17 and 25 (i.e., g_(k) ¹=ρ). Consequently, when the reception module fails to decode data of y_(k) ¹ correctly, it is possible to estimate the phase of g_(k) ¹ from the phase of y_(k) ¹. In other words, in this case, even though the CFO is estimated in a blind decoding manner, the phase ambiguity problem does not occur.

Meanwhile, after a case to which y_(k) ¹ belongs among the four cases is confirmed, the function z_(k) ^(n) can be generated through processing of y_(k) ^(n) as shown in Equation 13. Since all values of z_(k) ^(n) is in a state of in-phase (or co-phase), the final residual CFO can be estimated based on the generated z_(k) ^(n) and FIG. 14.

Meanwhile, according to the proposed embodiment, as |H_(k) ¹|² decreases and the residual CFO decreases, the accuracy of the residual CFO estimation can be improved because noise effects can be eliminated from the final residual CFO.

According to another proposed embodiment, the BPSK and the QPSK can be alternately used in a series of OFDM symbols. That is, similar to the case in which the BPSK scheme and the QBPSK scheme are alternately used, the BPSK and the QPSK can be alternately used in two OFDM symbols. However, in this case, since the phase of the product of two received signals is not placed at one of the four points shown in FIG. 24, the aforementioned method cannot be applied as it is.

In this case, the residual CFO can be estimated according to Equation 18.

$\begin{matrix} {{y_{k}^{n}\overset{\Delta}{=}{{\overset{\sim}{r}}_{k}^{n + 1}\left( {\overset{\sim}{r}}_{k}^{n} \right)}^{*}},{k \in C},{n = 1},2,\ldots\mspace{14mu},L} & \left\lbrack {{Equation}\mspace{14mu} 18} \right\rbrack \\ {{\overset{\sim}{r}}_{k}^{l} = \left\{ \begin{matrix} {r_{k}^{l}e^{j\frac{\pi}{4}}} & {{if}\mspace{14mu} B\; P\; S\; K} \\ r_{k}^{l} & {otherwise} \end{matrix} \right.} & \; \end{matrix}$

Equation 18 can be used instead of Equation 12. According to Equation 18, a phase of a signal received in the OFDM symbol where the BPSK is used can be uniformly changed by π/4. In this case, as shown in FIG. 26, a constellation of the BPSK of which the phase is changed partially matches that of the QPSK. Thus, the aforementioned embodiments can be equally applied to processes after Equation 18.

Meanwhile, in this embodiment, an angle for the phase change can be defined as

${\alpha\frac{\pi}{4}},\left( {{\alpha = {{2n} + 1}},{n \in {\mathbb{Z}}}} \right)$ (where Z is a set of integers). That is, it is meaningful that the BPSK constellation is changed as a part of the QPSK constellation and a changed phase value may be different. Moreover, according to the aforementioned embodiments, phase values of r_(k) ^(l) in all OFDM symbols where the BPSK is used can be uniformly changed by

${\alpha\frac{\pi}{4}},{\left( {{\alpha = {{2n} + 1}},{n \in {\mathbb{Z}}}} \right).}$

However, even when the phases of are r_(k) ^(l) rotated using different values, the result may be the same as the above result. For instance, when a phase value of r₁ ^(l) is rotated by π/4 and a phase value of r₂ ^(l) is rotated by 3π/4, the result may be the same as that of the case in which two received signals are rotated by the same phase value according to Equation 18. This is because points of the BPSK constellation are simply moved to points of the QPSK constellation.

The aforementioned embodiment can be equally applied to a case where the QBPSK and the QPSK are alternately used in each OFDM symbol. That is, when the QBPSK and the QPSK are used in two consecutive OFDM symbols, the aforementioned CFO estimation procedure can be equally applied by changing a phase value of a symbol where the QBPSK is used instead of the BPSK.

According to a further embodiment, when the BPSK and QPSK is used in a series of OFDM symbols, Equation 19 can be used instead of Equation 12.

$\begin{matrix} {{y_{k}^{n}\overset{\Delta}{=}{{r_{k}^{n + 1}\left( r_{k}^{n} \right)}^{*}e^{j\frac{\pi}{4}}}},{k \in C},{n = 1},2,\ldots\mspace{14mu},L} & \left\lbrack {{Equation}\mspace{14mu} 19} \right\rbrack \end{matrix}$

Unlike Equation 18, according to Equation 19, the entire phase of y_(k) ^(n) is changed by π/4 instead of changing phases of individual OFDM symbols. Equation 19 can be satisfied irrespective of whether the BPSK and the QPSK is used for an n^(th) OFDM symbol. This is because the BPSK and the QPSK are alternately used in every two consecutive OFDM symbols. Thus, it is possible to obtain the same result as that of Equation 18.

In Equation 19, even when phases of y_(k) ^(n) are rotated by different values, the same result occurs. For example, when a phase of y_(l) ^(n) is rotated by π/4 and a phase of y₂ ^(n) is rotated by 3π/4, the same result occurs. In addition, the embodiment described with reference to Equation 19 can be equally applied to not only the case where the BPSK and the QPSK are alternately used in every two OFDM symbols but also the case where the QBPSK and the QPSK are alternately used in every two OFDM symbols.

Hereinafter, a description will be given of UE's operation related to the aforementioned embodiments. First, while data transmitted from a transmission module is received at a reception module, a CFO occurring in an n^(th) OFDM symbol is defined as ϵ_(n). To eliminate the CFO in the n^(th) time-domain OFDM symbol and a CFO in an (n+1)^(th) time-domain OFDM symbol, the reception module estimates CFO(s) using a preamble part of a firstly received frame. The estimated CFO is defined as {circumflex over (ϵ)}_(n). However, since the primarily estimated CFO is not complete (i.e., ϵ_(n)≠{circumflex over (ϵ)}_(n)), there must be a residual CFO (Δ{circumflex over (ϵ)}_(n)). Thus, the reception module can estimate the residual CFO using the aforementioned embodiments independently or any combination thereof.

After estimating the residual CFO, the reception module can correct a phase of a signal r_(k) ^(l) received in a subcarrier k as shown in Equation 20 to eliminate the effect of the estimated residual CFO from the received signal. {tilde over (r)} _(k) ^(l) =r _(k) ^(l) e ^(−j2πΔ{tilde over (ϵ)}(N+N) ^(g) ^()/N) ,l=n,n+1  [Equation 20]

By doing so, the reception module can compensate phase distortion in the received signal due to the residual CFO and thus, reception SINR can finally be improved.

Meanwhile, unlike a method of eliminating a CFO in the time domain, according to the method of eliminating a CFO in the frequency domain, it is impossible to cancel the effect of a leak signal which occurs due to the CFO. Therefore, to eliminate the CFO effects from (n+2)^(th) and (n+3)^(th) time-domain OFDM symbols, a CFO estimation value shown in Equation 21 can be used. {circumflex over (ϵ)}_(n+2)={circumflex over (ϵ)}_(n)+Δ{circumflex over (ϵ)}_(n)  [Equation 21]

In Equation 21, since {circumflex over (ϵ)}_(n+2) is closer to ϵ_(n+2) than {circumflex over (ϵ)}_(n), there may be a smaller residual CFO. Thus, a frequency-domain received signal may have less signal leakage after FFT operation and thus efficiency can be improved in terms of a reception SNR. Next, Δ{circumflex over (ϵ)}_(n+2) is estimated using the proposed embodiments and then the phase of the signal received in the subcarrier can be corrected as described in Equation 20.

FIG. 27 is a flowchart illustrating the CFO estimation method according to the proposed embodiment. Specifically, FIG. 27 shows a time series flow of the CFO estimation method according to the embodiments described with reference to FIGS. 24 to 26. Thus, it is apparent that although the aforementioned features described with reference to FIGS. 18 to 22 are not explicitly shown and described in FIG. 23, the features can be applied to the flowchart in FIG. 23 in the same or similar manner Thus, it is apparent that although the aforementioned features described with reference to FIGS. 24 to 26 are not explicitly shown and described in FIG. 27, the features can be applied in the same or similar manner.

First, a transmission module transmits data to a reception module S2710. In this case, data is transmitted in a unit of frame, which is defined by an OFDM symbol and a subcarrier. In addition, the data is mapped to a specific constellation and then transmitted to the reception module. As such as a constellation, the QPSK can be used. Moreover, the BPSK (or QBPSK) and the QPSK can be alternately used in a series of consecutive OFDM symbols.

Meanwhile, the reception module primarily estimate a CFO value based on a received signal [S2720]. Such a process is performed using a preamble part such as an L-STF, an L-LTF, and the like. However, since a CFO is changed depending on time, the CFO value estimated in the step S2720 may be inaccurate.

Therefore, the reception module estimates a residual CFO value to compensate the primarily estimated CFO value [S2730]. As described above, the reception module estimates the residual CFO value on the assumption that channels of received signals in two consecutive OFDM symbols are equal to each other. Specifically, the reception module calculate a product of the two received signal and then checks signs and magnitudes of a real part and an imaginary part of the product based on the assumption that the residual CFO value is smaller than the primarily estimated CFO value. If signals received in the total subcarriers are in a state of in-phase, the reception module can obtain the residual CFO from a phase value calculated by adding results of all the subcarriers.

Finally, the reception module can accurately decode the data transmitted from the transmission module by eliminating effects of the CFO estimated in the step S2720 and the residual CFO estimated in the step S2730 from the received signal.

5. Third Proposed Method for CFO Estimation

In the foregoing description, a method for a reception module to estimate a CFO has been explained when data transmission is performed using BPSK (QBPSK) or QPSK. In this case, the reception module efficiently operates when a residual CFO has a size relatively smaller than a size of a firstly estimated CFO. However, if the residual CFO is estimated using a blind type, the reception module is unable to guarantee that the size of the residual CFO is sufficiently small. Hence, when the size of the residual CFO is large, a method for the reception module to estimate a CFO is explained in the following.

In FIGS. 28 and 29 described in the following, a method for a reception module to estimate a CFO is explained by 4 steps. FIG. 28 is a diagram illustrating a resource block for explaining a proposed embodiment. FIG. 28 shows a resource block (RB) consisting of a plurality of resource elements (REs) defined in LTE (long term evolution)/LTE-A (LTE-advanced). Referring to FIG. 28, one RB is defined by 14 OFDM symbols arranged on a horizontal axis and 12 subcarriers arranged on a vertical axis and the RB includes 168 REs in total.

In FIG. 28, REs expressed by slashes correspond to RSs (reference signals). In this case, REs expressed by slashes slashed from the top left to the bottom right correspond to CRSs (cell-specific RSs) and REs expressed by slashes slashed from the top right to the bottom left correspond to DMRSs (demodulation RSs). Values of the RSs are known to a reception module in advance and the reception module determines CSI (channel state information) or performs channel estimation for performing demodulation using an RS.

According to a proposed embodiment, a reception module estimates a first residual CFO using an RS (hereinafter, a first residual CFO). The reception module compensates for a received data using the estimated first residual CFO. In particular, when the reception module is unable to know a size of the total residual CFO, the reception module controls a size of the residual CFO to be included within a small range. Subsequently, the reception module estimates the residual CFO according to the aforementioned first CFO estimation method or the second CFO estimation method (hereinafter, a second residual CFO). In other word, in order for the reception module to efficiently apply the blind type CFO estimation method, it is necessary to control the residual CFO with a small size. Hence, the first residual CFO is estimated using embodiments described in the following. The total residual CFO (hereinafter, a third residual CFO) corresponds to the sum of the first residual CFO and the second residual CFO. In the following, the embodiments proposed in the present invention are explained in detail.

First of all, a reception module estimates a first residual CFO using an RS. Similar to the aforementioned first CFO estimation method and the second CFO estimation method, the reception module generates a function y_(k) ^(n) of equation 22 using signals received on two consecutive OFDM symbols. y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*,k∈D _(n) ,n=1,2, . . . ,L  [Equation 22]

A unique point of equation 22 different from equations 5 and 12 is in that k corresponds to an element of D_(n) rather than an element of C_(n). The D_(n) corresponds to a set indicating an index of a subcarrier at which an RS is positioned in an n^(th) OFDM symbol. In particular, the reception module generates the y_(k) ^(n) via the equation 22 using a relation between an RE at which the RS is located and consecutive OFDM symbols. Meanwhile, the C_(n) corresponds to a set indicating an index of a subcarrier at which data rather than the RS is located. Since either an RS or data is transmitted only in a manner of being loaded on a single RE, it satisfies {D∩C}={ϕ}.

In addition to the equation 22, equation 23 in the following describes a procedure that the reception module measures the first residual CFO.

$\begin{matrix} {{\Delta\;{\hat{\epsilon}}_{n}^{(1)}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in D_{n}}y_{k}^{n}}} \right)} \times \frac{N}{2\;{\pi\left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 23} \right\rbrack \end{matrix}$

In equation 23, Δ{circumflex over (ϵ)}_(n) ⁽¹⁾ corresponds to a first residual CFO measured for an n^(th) OFDM symbol using an RS. In particular, the first residual CFO corresponds to a CFO measured for only a subcarrier D_(n) at which an RS is located.

If the first residual CFO is determined by the equations 22 and 23, the reception module compensates for data received using the first residual CFO. The abovementioned procedure can also be comprehended as a procedure of changing phases of the data as much as the calculated first residual CFO. Since the RS corresponds to a value known to the reception module in advance, the first residual CFO calculated via the equations 22 and 23 can compensate for the total CFO having a large value with a sufficiently small value. Although the first residual CFO is not completely identical to the total CFO, the remaining residual CFO can be reduced to a sufficiently small value via the procedure of compensating for data using the first residual CFO.

Meanwhile, if the equation 23 is explained in detail, the y_(k) ^(n) can be represented as equation 24 described in the following.

$\begin{matrix} \begin{matrix} {y_{k}^{n} \approx {{H_{k}^{n}}^{2}{s_{k}^{n + 1}\left( s_{k}^{n} \right)}^{*}e^{j\; 2\;\pi\;\Delta\;{{\epsilon_{n}{({N + N_{g}})}}/N}}}} \\ \left. \rightarrow{{H_{k}^{n}}^{2}\rho\; e^{j\; 2\;\pi\;\Delta\;{{\epsilon_{n}{({N + N_{g}})}}/N}}} \right. \end{matrix} & \left\lbrack {{Equation}\mspace{14mu} 24} \right\rbrack \end{matrix}$

A first part of the equation 24 is derived from the equation 8 and the equation 9. Meanwhile, as mentioned in the foregoing description, since an RS corresponds to a value known to the reception module in advance, the reception module knows a phase of s_(k) ^(n+1)(s_(k) ^(n))* in advance. Hence, if a component of the s_(k) ^(n+1)(s_(k) ^(n))* is eliminated from the equation 24, a second part of the equation 24 is obtained. Since information on a phase value of the s_(k) ^(n+1)(s_(k) ^(n))* component is known to the reception module in advance, it is not necessary to have a procedure for solving a phase ambiguity problem in the equations 22 and 23.

Meanwhile, as mentioned in the foregoing description, if the first residual CFO is determined, a procedure of compensating for data is represented as equation 25 in the following. y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*e ^(−j2πΔ{tilde over (ϵ)}) ^(n) ⁽¹⁾ ^((N+N) ^(g) ^()/N) ,k∈D _(n) ,n=1,2, . . . ,L  [Equation 25]

The equation 25 corresponds to a procedure for changing a phase of a reception signal as much as the first residual CFO (Δ{circumflex over (ϵ)}_(n) ⁽¹⁾) measured via the equation 23. If the data compensation procedure using the first residual CFO is completed, the reception module estimates a second residual CFO (Δ{circumflex over (ϵ)}_(n) ⁽²⁾) among the total residual CFO (Δ{circumflex over (ϵ)}_(n)) using the aforementioned first CFO estimation method or the second CFO estimation method. When data transmission is performed using QPSK, the procedure of estimating the second residual CFO is represented as equation 26 described in the following.

$\begin{matrix} {z_{k}^{n} = \left\{ \begin{matrix} y_{k}^{n} & {{{{if}\mspace{14mu}{{real}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\ {- y_{k}^{n}} & {{{{if}\mspace{14mu}{{real}\left( y_{k}^{n} \right)}} < 0},{{{real}\left( y_{k}^{n} \right)} \geq {{imag}\left( y_{k}^{n} \right)}}} \\ {{- j} \times y_{k}^{n}} & {{{{if}\mspace{14mu}{{imag}\left( y_{k}^{n} \right)}} \geq 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}} \\ {j \times y_{k}^{n}} & {{{{if}\mspace{14mu}{{imag}\left( y_{k}^{n} \right)}} < 0},{{{real}\left( y_{k}^{n} \right)} < {{imag}\left( y_{k}^{n} \right)}}} \end{matrix} \right.} & \left\lbrack {{Equation}\mspace{14mu} 26} \right\rbrack \end{matrix}$

The equation 26 is identical to the equation 13 and indicates a procedure for changing a phase of the total data into an in-phase state to estimate the second residual CFO which has reduced to a small size. If BPSK or QPBSK is used instead of QPSK, the equation 6 is used for the same procedure. Meanwhile, if BPSK and QBPSK are alternately used for two consecutive OFDM symbols, it may use the equation 11 instead of the equation 25. Or, if BPSK (or, QBPSK) and QPSK are alternately used for two consecutive OFDM symbols, it may use the equation 18 instead of the equation 25.

In addition to the equation 26, the second residual CFO is estimated according to equation 27.

$\begin{matrix} {{\Delta\;{\hat{\epsilon}}_{n}^{(2)}} = {{angle}\left\{ {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in C_{n}}z_{k}^{n}}} \right) \times \frac{N}{2\;{\pi\left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 27} \right\rbrack \end{matrix}$

Unlike the equation 23, the equation 27 calculates a residual CFO for C_(n). In particular, a residual CFO is calculated for subcarriers at which data exists only except a subcarrier at which an RS exists. If the second residual CFO is finally calculated via the equation 27, the total CFO can be represented by the sum of the first residual CFO and the second residual CFO (equation 28). Δ{circumflex over (ϵ)}_(n)=Δ{circumflex over (ϵ)}_(n) ⁽¹⁾+Δ{circumflex over (ϵ)}_(n) ⁽²⁾  [Equation 28]

In the following, various embodiments capable of being applied to a procedure for estimating a CFO are explained according to a series of aforementioned procedures.

According to one embodiment, a reception module can calculate a residual CFO not only for a subcarrier at which data exists bit also for a subcarrier at which an RS exists in the course of performing a procedure of estimating a second residual CFO. In particular, the second residual CFO can be calculated via equation 29 rather than the equation 27.

$\begin{matrix} {{\Delta\;{\hat{\epsilon}}_{n}^{(2)}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}\left( {{\sum\limits_{k \in C_{n}}z_{k}^{n}} + {\sum\limits_{k \in D_{n}}{y_{k}^{n}e^{{- j}\; 2\;\pi\;\Delta\;{{{\hat{\epsilon}}_{n}^{(1)}{({N + N_{g}})}}/N}}}}} \right)} \right)} \times \frac{N}{2\;{\pi\left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 29} \right\rbrack \end{matrix}$

According to the abovementioned embodiment, since the number of samples used for estimating the second residual CFO is increased as many as the number of subcarriers at which an RS exists, performance of estimating the second residual CFO is enhanced. For example, if the number of data subcarriers is identical to the number of RS subcarriers (|C|=|D|), performance for estimating the second residual CFO according to the equation 29 is better than performance for estimating the second residual CFO according to the equation 27 as much as 3 dB. Moreover, since the second residual CFO is defined in a form of improving the accuracy of the first residual CFO, performance for estimating the total residual CFO (or, a third residual CFO) is also enhanced as much as 3 dB.

In the foregoing description, if the first residual CFO is calculated, the reception module compensates for data using the calculated first residual CFO and then calculates the second residual CFO. On the contrary, according to one embodiment, a procedure of compensating for data and a procedure of calculating the second residual CFO can be performed by a single procedure in a manner of being integrated according to equation 30.

$\begin{matrix} {{\Delta\;{\hat{\epsilon}}_{n}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}\left( {{\sum\limits_{k \in C_{n}}{z_{k}^{n}e^{j\; 2\;\pi\;\Delta\;{{{\hat{\epsilon}}_{n}^{(1)}{({N + N_{g}})}}/N}}}} + {\sum\limits_{k \in D_{n}}y_{k}^{n}}} \right)} \right)} \times \frac{N}{2\;{\pi\left( {N + N_{g}} \right)}}}} & \left\lbrack {{Equation}\mspace{14mu} 30} \right\rbrack \end{matrix}$

Specifically, if the first residual CFO is calculated, a third residual CFO can be immediately calculated without the procedure of compensating for data and the procedure of calculating the second residual CFO. To this end, a phase of a reception signal (z_(k) ^(n)) is changed as much as the first residual CFO in the equation 30. And, according to the present embodiment, similar to the equation 29, the third residual CFO is calculated using both a data carrier and an RS carrier. The equation 30 can be represented as equation 31 via mathematical calculation. By doing so, it is able to know that the total residual CFO calculated via the equation 30 is identical to the total residual CFO according to the equations 22 to 28.

[Equation  31] $\begin{matrix} {{\Delta\;{\hat{\epsilon}}_{n}} = {{\Delta\;{\hat{\epsilon}}_{n}^{(1)}} + {{{angle}\left( {\sum\limits_{n = 1}^{L}\left( {{\sum\limits_{k \in C_{n}}z_{k}^{n}} + {\sum\limits_{k \in D_{n}}{y_{k}^{n}e^{{- j}\; 2\;\pi\;\Delta\;{{{\hat{\epsilon}}_{n}^{(1)}{({N + N_{g}})}}/N}}}}} \right)} \right)} \times}}} \\ {\frac{N}{2\;\pi\;\left( {N + N_{g}} \right)}} \\ {= {{\Delta\;{\hat{\epsilon}}_{n}^{(1)}} + {\Delta\;{\hat{\epsilon}}_{n}^{(2)}}}} \end{matrix}$

In the foregoing description, a CFO estimation method has been explained under a condition (D_(n)=D_(n+1)) that a subcarrier at which an RS is positioned is identical to each other in two consecutive OFDM symbols. However, the condition may not be satisfied depending on a form of deploying data and an RS to a resource region. For example, it may consider such a case as D_(n)⊂D_(n+1) or D_(n+1)⊃D_(n).

In particular, when a reception module performs a procedure of estimating the first residual CFO via an RS, the reception module can perform the procedure on a subcarrier only where an RS exists in both an n^(th) OFDM symbol and an (n+1)^(th) OFDM symbol. This is because, if the reception module knows either s_(k) ^(n) or s_(k) ^(n+1) only, a phase ambiguity problem occurs.

Moreover, if the first residual CFO is estimated according to a condition of D _(n)={D_(n)∩D_(n+1)}, C_(n) corresponding to a set of subcarriers at which data exists in an n^(th) OFDM symbol can also be defined as C _(n)=C_(n)∪{D_(n)−{D_(n)∩D_(n+1)}}. In particular, when an RS exists in both two contiguous OFDM symbols in a subcarrier, subcarriers except the subcarrier are excluded from the procedure of estimating the first residual CFO and the subcarriers are used as samples in the procedure of estimating the first residual CFO.

In the foregoing description, a CFO estimation method has been explained on the basis of a subcarrier of which an RS is positioned at two consecutive OFDM symbols. Yet, if a subcarrier of which an RS is positioned at two consecutive OFDM symbols does not exist, such a case as D _(n)={ϕ} may occur. In this case, the aforementioned procedure for estimating the first residual CFO can be omitted. In particular, a following procedure can be performed by setting a first residual CFO (Δ{circumflex over (ϵ)}_(n) ⁽¹⁾) to 0.

On the contrary, the aforementioned first residual CFO estimation procedure can also be applied to OFDM symbols away from each other as much as G. In particular, as shown in equation 32, it may be able to calculate the first residual CFO for a subcarrier of which an RS exists in both an n^(th) OFDM symbol and an (n+G)^(th) OFDM symbol.

$\begin{matrix} {{{y_{k}^{n}\overset{\Delta}{=}{r_{k}^{n + G}\left( r_{k}^{n} \right)}^{*}},{k \in {\overset{\_}{D}}_{n,G}},{n = 1},2,\ldots\mspace{14mu},L}{{\Delta\;{\hat{\epsilon}}_{n}^{(1)}} = {{{angle}\left( {\sum\limits_{n = 1}^{L}{\sum\limits_{k \in D_{n}}{y_{k}^{n}\left( s_{k}^{n + G} \right)}^{*}}} \right)} \times \frac{N}{2\;\pi\;{G\left( {N + N_{g}} \right)}}}}} & \left\lbrack {{Equation}\mspace{14mu} 32} \right\rbrack \end{matrix}$

In the equation 32, D _(n,G)≙{D_(n)∩D_(n+G)} is defined. In other word, in the foregoing description, a CFO estimation method has been explained under a condition that a channel and a CFO do not change in two contiguous OFDM symbols. On the contrary, referring to the equation 32, a reception module estimates a CFO under a condition that a channel and a CFO do not change during G+1 number of OFDM symbols. An error e_(k) ^(n) according to the abovementioned procedure can be defined as equation 33 in the following. e _(k) ^(n) ≙|y _(k) ^(n) −|H _(k) ^(n)|² s _(k) ^(n+G)(s _(k) ^(n))*e ^(j2πΔϵ) ^(n) ^((N+N) ^(g) ^()G/N)|²  [Equation 33]

The error e_(k) ^(n) increases as a distance (G) between two OFDM symbols is getting bigger. Hence, in the present embodiment, if D _(n,G)≙{D_(n)∩D_(n+G)}≠{ϕ} is satisfied and a channel and a CFO do not considerably change during G+1 number of OFDM symbols, efficiency of estimating a CFO can be increased.

In the foregoing description, a CFO is estimated on a subcarrier common to two OFDM symbols (two contiguous OFDM symbols or OFDM symbols away from each other as much as G). Yet, according to a different embodiment, a CFO can also be estimated via contiguous subcarriers rather than the same subcarrier. For example, a procedure for estimating the first residual CFO can be changed like equation 34 described in the following. y _(k) ^(n) ≙r _(k+δ) _(D) ^(n+1)(r _(k) ^(n))*,k∈D _(n) ,n=1,2, . . . ,L  [Equation 34]

In the equation 34, δ_(D) corresponds to a distance between two subcarriers used for estimating a CFO. If the first residual CFO is calculated based on the equation 34, a procedure for calculating the second residual CFO can be changed like equation 35 described in the following. y _(k) ^(n) ≙r _(k+δ) _(C) ^(n+1)(r _(k) ^(n))*e ^(−j2πΔ{tilde over (ϵ)}) ^(n) ⁽¹⁾ ^((N+N) ^(g) ^()/N) ,k∈D _(n) ,n=1,2, . . . ,L  [Equation 35]

In the equation 35, δ_(C) also corresponds to a distance between adjacent subcarriers. In the equations 34 and 35, assume that channels between adjacent subcarriers are identical to each other (H_(k+δ) ^(n+1)=H_(k) ^(n)). An error caused by the assumption is represented by equation 36 in a manner of being defined by an equation similar to the equation 33. e _(k) ^(n) ≙|y _(k) ^(n) −|H _(k) ^(n)|² s _(k) ^(n+1)(s _(k) ^(n))*e ^(−j2πΔϵ) ^(n) ^((N+N) ^(g) ^()/N)|²  [Equation 36]

where H_(k+δ) ^(n+1)(H_(k) ^(n))*≈|H_(k) ^(n)|²

An error e_(k) ^(n) increases as a difference between H_(k) ^(n) and H_(k+δ) ^(n+1) is getting bigger. Hence, in the present embodiment, as selectivity of a frequency axis is getting smaller, efficiency is increasing. For example, if a delay profile of a channel is small similar to in-door environment, the abovementioned embodiment efficiently works.

Meanwhile, in the equations 34 and 35, the δ_(C) and the δ_(D) can be configured by a different value. If the two values are configured by 0, the equations 34 and 35 are identical to the equations 22 and 25.

According to the third proposed method for CFO estimation, a reception module estimates a first residual CFO using an RS known to the reception module in advance. The first residual CFO is used for a procedure of compensating for data. A second residual CFO is calculated from the compensated data according to the first CFO estimation method or the second CFO estimation method.

Meanwhile, the procedure for estimating the first residual CFO using an RS has been explained under the condition that a channel is common to two OFDM symbols. Yet, in case of Case A shown in FIG. 28, CRSs are transmitted via a different antenna port in a first OFDM symbol and a second OFDM symbol. In particular, since channels are different from each other in the two OFDM symbols, it is unable to apply the abovementioned scheme as it is. Yet, in case of E-PDCCH (evolved physical downlink control channel) in LTE/LTE-A, since two OFDM symbols are transmitted via the same antenna port, it may be able to apply the proposed CFO estimation method as it is.

Referring to FIG. 28, an RS is not assigned to a 3^(rd) OFDM symbol and a 4^(th) OFDM symbol. Hence, the total residual CFO is estimated without a procedure for estimating the first residual CFO. Among the 4^(th) OFDM symbol and a 5^(th) OFDM symbol, since an RS exists at the 5^(th) OFDM symbol only, D _(n)={ϕ} is satisfied. Hence, the procedure for estimating the first residual CFO is omitted.

Meanwhile, in case of Case B shown in FIG. 28 (6^(th) and 7^(th) OFDM symbols), RSs twice as many as RSs of the Case A are assigned. In particular, since the Case B has double samples for estimating the first residual CFO, performance of estimating the first residual CFO is improved as much as 3 dB compared to the Case A.

In case of Case C (9^(th) and 11^(th) OFDM symbols), it may be able to estimate the first residual CFO according to the equation 33 on the basis of OFDM symbols away from each other as much as G=1. Yet, it is highly probable that estimation performance is to be deteriorated compared to the Cases A and B.

According to the aforementioned embodiments, a reception module estimates the first residual CFO using data known to the reception module in advance to compensate for a part of CFOs of the data (a phase ambiguity problem does not occur in this procedure) and estimates the second residual CFO according to the aforementioned blind type CFO estimation method. Since it is able to apply the blind type CFO estimation method to a case that a size of a residual CFO is small, performance and efficiency in estimating a residual CFO can be enhanced.

FIG. 29 is a flowchart for a method of estimating a CFO according to the present invention.

FIG. 29 illustrates the CFO estimation method according to the embodiments mentioned earlier in FIG. 28 according to a time serial flow. Hence, although the embodiments mentioned earlier in FIG. 28 are not explained in detail in FIG. 29, the embodiments can be similarly or identically applied.

First of all, a transmission module transmits data to a reception module [S2910]. The data can be transmitted in a unit of a frame defined by an OFDM symbol and a subcarrier. The data is transmitted to the reception module in a manner of being mapped to a specific constellation. The constellation may use BPSK, QBPSK, QPSK, etc. BPSK and QBPSK can be alternately used for a series of contiguous (or consecutive) OFDM symbols or BPSK (or, QBPSK) and QPSK can be alternately used for the contiguous OFDM symbols.

Meanwhile, the reception module firstly estimates a CFO value from a reception signal [S2920]. This procedure is performed using such a preamble part as L-STF, L-LTF, and the like of a frame. Yet, since the CFO value changes over time, the CFO value estimated in the step S2920 may not be an accurate CFO value. Hence, the reception module performs a procedure for estimating a residual CFO in succession.

The reception module estimates a first residual CFO value using an RS [S2930]. A procedure for estimating the first residual CFO value is performed by measuring the first residual CFO using RSs received from two contiguous OFDM symbols on a specific subcarrier. The reception module compensates for a phase value of data using the first residual CFO and estimates a second residual CFO on the basis of the compensated data [S2940]. Since the data is compensated via the first residual CFO estimated in the step S2930, the second residual CFO estimated in the step S2940 may have a relatively small value. In particular, the reception module estimates the second residual CFO according to the step S2330 of FIG. 23 and the step S2730 of FIG. 27. The sum of the two residual CFOs estimated in the steps S2930 and S2940 becomes the total residual CFO.

Lastly, the reception module is able to precisely decode the data transmitted by the transmission module by eliminating the impact of the CFO estimated in the step S2920 and the total residual CFO estimated in the steps S2930 and S2940.

6. Fourth Proposed Method for CFO Estimation

In the foregoing description, embodiments for a reception module to calculate a residual CFO using the first, the second and the third CFO estimation method have been explained. Yet, since the embodiments explain a case of transmitting data using BPSK, QBPSK, and QPSK modulation scheme only, it is unable to apply the embodiments to 16-QAM (16-Quadrature Amplitude modulation) corresponding to a modulation scheme of a higher order, and the like. This is because the aforementioned blind type CFO estimation method is performed under a condition that a phase difference between signals received from two contiguous OFDM symbols on a specific subcarrier is gathered to a specific value. If a modulation scheme of an order equal to or higher than an order of the 16-QAM modulation scheme is used, the condition is not valid anymore.

Hence, when data is transmitted using a modulation scheme of a high order (e.g., 16-QAM), a method for a reception module to estimate a CFO is explained in the following with reference to FIGS. 30 to 32.

First of all, as mentioned earlier in the proposed embodiments, a pair of received signals corresponds to signals received from two contiguous OFDM symbols. In the equation 5, C indicates a set of subcarrier indexes of the total reception signal pairs and the aforementioned CFO estimation performance is proportional to a size of the set C. In particular, as the number of samples (reception signal pairs) applied to a CFO estimation algorithm is getting bigger, the estimation performance is more enhanced. As mentioned in the foregoing description, if data transmission is performed using BPSK, QBPSK, and QPSK, all reception signal pairs are used for estimating a CFO.

Meanwhile, if the CFO estimation method is applied to a high order QAM (e.g., 16-QAM), only a part of the total reception signal pairs can be used as a sample. Hence, the equation 5 can be represented as equation 37 descried in the following. y _(k) ^(n) ≙r _(k) ^(n+1)(r _(k) ^(n))*,k∈C,n=1,2, . . . ,L[Equation 37]

where C⊂C

In the equation 37, C corresponds to a subset selected from the C corresponding to the set of total reception signal pairs via an embodiment described in the following. In particular, an object of the proposed embodiment is to maximize performance by increasing the number of samples for estimating a CFO through a method of selecting a reception signal pair for maximizing a size of the C.

FIGS. 30 and 31 are diagrams for a method of segmenting 16-QAM constellation according to embodiments proposed in the present invention. As shown in FIGS. 30 and 31, a part of points located on 16-QAM constellation is matched with points located on QPSK constellation. Hence, if signals received from two OFDM symbols are transmitted in a manner of being mapped to points corresponding to QPSK among the points located on the 16-QAM constellation, it may be able to apply the first, the second, and the third CFO estimation methods as it is. Based on this, a method of estimating a CFO using a size and a phase of a reception signal pair is explained in the following.

First of all, according to the proposed embodiment, a reception module compares sizes between a pair of received signals and determines whether to use the pair of received signals in a procedure for estimating a residual CFO. When the reception module compares the sizes between the pair of received signals, the reception module can calculate at least one of a ratio of the sizes and a difference between the sizes. Equation 38 illustrates a procedure of comparing the ratio of the sizes of the pair of received signals and equation 39 illustrates a procedure of comparing the difference between the sizes of the pair of received signals.

$\begin{matrix} {\gamma_{k}^{n}\overset{\Delta}{=}{\frac{r_{k}^{n + 1}}{r_{k}^{n}}\mspace{14mu}{or}\mspace{14mu}{\frac{r_{k}^{n + 1}}{r_{k}^{n}}}}} & \left\lbrack {{Equation}\mspace{14mu} 38} \right\rbrack \\ {\kappa_{k}^{n}\overset{\Delta}{=}{{r_{k}^{n + 1}} - {{r_{k}^{n}}\mspace{14mu}{or}\mspace{14mu}{{r_{k}^{n + 1} - r_{k}^{n}}}}}} & \left\lbrack {{Equation}\mspace{14mu} 39} \right\rbrack \end{matrix}$

In equations 38 and 39, r_(k) ^(n) corresponds to a signal received from an n^(th) OFDM symbol and k indicates subcarrier indexes at which the n^(th) OFDM symbol and an (n+1)^(th) OFDM symbol are located. In the equations 38 and 39, (r_(k) ^(n), r_(k) ^(n+1)) corresponds to the aforementioned pair of received signals.

Meanwhile, in consideration of a condition that two channels on the same subcarrier are identical to each other in the n^(th) OFDM symbol and the (n+1)^(th) OFDM symbol and a CFO and noise do not exist in the n^(th) OFDM symbol and the (n+1)^(th) OFDM symbol, the equation 38 can be represented as equation 40 described in the following.

$\begin{matrix} {\gamma_{k}^{n}\overset{\Delta}{=}{\frac{r_{k}^{n + 1}}{r_{k}^{n}} = {\frac{{H_{k}^{n}s_{k}^{n + 1}}}{{H_{k}^{n}s_{k}^{n}}} = {\frac{{H_{k}^{n}}\;{s_{k}^{n + 1}}}{{H_{k}^{n}}\;{s_{k}^{n}}} = \frac{s_{k}^{n + 1}}{s_{k}^{n}}}}}} & \left\lbrack {{Equation}\mspace{14mu} 40} \right\rbrack \end{matrix}$

In particular, since a channel is the same in two OFDM symbols, a size ratio of a pair of received signals received by a reception module can be represented by a size ratio of data symbols s_(k) ^(n),s_(k) ^(n+1) transmitted by a transmission module. If sizes of data symbols transmitted by the transmission module are the same, a size ratio value of a pair of received signals becomes 1.

FIG. 30 illustrates an embodiment of segmenting 16-QAM constellation into 3. In particular, if (A), (b), and (C) shown in FIG. 30 are summed up, it becomes 16-QAM constellation. In particular, 16-QAM constellation is segmented into (A), (b), and (C) according to a data symbol having the same power. Hence, if a size ratio γ_(k) ^(n) of a pair of signals received from a reception module corresponds to 1, it is able to know that two data symbols constructing the pair of the received signals belong to at least one of (A), (b), and (c).

For example, if the size ratio of the pair of the received signals corresponds to 1, the two data symbols may correspond to one selected from the group consisting of (a1, a2), (a1, a4), (b1, b2), and (c1, c3). However, the two data symbols are unable to become (a1, b1) or (b1, c1). Consequently, a reception module is able to find out a pair of received signals corresponding to two data symbols having the same power using a size ratio of the pair of the received signals without using channel information (i.e., using a non-coherent scheme).

Meanwhile, in FIG. 30, (b) and (c) have an arrangement form identical to QPSK constellation, respectively. Hence, if it is determined that the pair of the received signals exists in one of (b) and (c) at the same time, the reception module is able to estimate a residual CFO using the aforementioned second CFO estimation method.

On the contrary, the pair of the received signals may belong to (A). A phase difference between data symbols can be classified into two types in the (A) group. First of all, a phase difference between points expressed by the same marker (quadrangle or circle) corresponds to one selected from the group consisting of {0, π/2, π, 3π/2}. And, a phase difference between points expressed by a different marker corresponds to one selected from the group consisting of {46°, 136°, 226°, 316°}.

In case of the first case (points expressed by the same marker), similar to (B) and (C), since a phase difference between symbols is identical to that of QPSK, it may be able to similarly apply the second CFO estimation method (hereinafter, (A-1) case). On the contrary, in case of the second case (points expressed by a different marker), since a phase difference is different between received signals, it is unable to apply the aforementioned CFO estimation methods as it is (hereinafter, (A-2) case). In case of the (A-1), the (b), and the (C), a reception module is able to immediately estimate a residual CFO through the second CFO estimation method. However, in case of the (A-2) case, if the reception module changes a phase of received signals as much as −46°, the reception module can process the (A-2) case in a manner of being identical to the case of the (A-1) case. In particular, it is necessary for the reception module to distinguish the (A-1) case from the (A-2) case. Regarding this, it shall be described later.

FIG. 30 illustrates a different embodiment of segmenting 16-QAM constellation into 2. Similar to FIG. 30, if (A) and (B) shown in FIG. 31 are summed up, it becomes 16-QAM constellation. If a size ratio of a pair of received signals corresponds to ⅓ or 3, a size difference between two data symbols selected by a transmission module has a difference as much as three times. In this case, the data symbols of the pair of the received signals belong to FIG. 31 (b) at the same time. FIG. 31 (b) corresponds to the sum of FIGS. 30 (b) and (c). Hence, although a size ratio of a pair of received signals corresponds to ⅓ or 3, a reception module is able to estimate a CFO by applying the second CFO estimation method. In other word, when a size ratio of a pair of received signals corresponds to ⅓ or 3, if the reception module applies the second CFO estimation method, the number of samples used for estimating a CFO increases (i.e., a size of C increases) compared to a case of applying the second CFO estimation method only when a size ratio corresponds to 1, thereby enhancing CFO estimation performance. For example, although a size ratio of a pair of received signals corresponds to ⅓ or 3, if the reception module allows the ratio to be used as a sample for a method of estimating a CFO, the reception module is able to estimate a CFO using such a pair of received signals as (b1, c2) and (b2, c2) shown in FIG. 31 (B). (Yet, such a pair of received signals as (a1, b1) cannot be used as a CFO estimation candidate.)

The aforementioned embodiment is summarized in the following. If a size ratio of a pair of received signals belongs to a prescribed range, a reception module uses the pair of received signals in the course of performing a procedure for estimating a residual CFO. According to one embodiment, a pair of received signals can be selected on the basis of a size difference rather than the size ratio of the pair of received signals.

In particular, theoretically, although it is accurate to determine whether or not a size ratio of a pair of received signals is identical to a specific value, when a system is actually implemented, channels on which the pair of received signals is transmitted are not the perfectly same. And, an impact of noise and a CFO exist all the time. Hence, it is difficult to apply the propose embodiment only when the size ratio of the pair of received signals is perfectly matched with a prescribed value. Hence, if it is determined that the size ratio has a value belonging to a prescribed range of the prescribed value, it may apply the proposed embodiment. In particular, if a size ratio of a pair of received signals satisfies a range of equation 41, the pair of received signals can be used in the course of performing a procedure for estimating a CFO. 1−δ≤γ_(k) ^(n)≤1+δ or 3−δ≤γ_(k) ^(n)≤3+δ  [Equation 41]

In the equation 41, δ corresponds to a threshold range for which a size ratio of a pair of received signals is allowed. For example, the δ can be configured by 0.1.

In the following, an embodiment for distinguishing the (A-1) case from the (A-2) case is additionally explained. As mentioned earlier in FIG. 30, if a size ratio of a pair of received signals corresponds to 1, (b) and (C) can be utilized as samples for performing the second CFO estimation method. Not only the (A-1) case but also the (A-2) case requires a procedure of changing phases of received signals. Hence, a method of identifying the (A-2) case is explained in the following.

According to an embodiment proposed in the present invention, a reception module can determine whether to use a pair of received signals as a sample for performing a method of estimating a residual CFO in consideration of a phase difference of the pair of received signals. A method of calculating a phase of a pair of received signals received by a reception module using a phase of a data symbol transmitted by a transmission module is explained before the embodiment is explained. In the aforementioned second CFO estimation method, a phase of y_(k) ^(n) corresponds to a phase of multiplication of a pair of signals received by a reception module. In this case, the phase is identical to a phase of multiplication of data symbols transmitted by a transmission module. This is because, when the pair of received signals is multiplied, phases of channels are cancelled out. Hence, it is able to know that a phase of a channel experienced by a reception signal does not influence on the phase of multiplication of the pair of received signals. In particular, although the phase of multiplication of the pair of received signals is replaced with the phase of multiplication of the data symbols transmitted by the transmission module irrespective of a channel of a received signal, there is no problem.

Meanwhile, in order to identify the (A-2) case of FIG. 30, the reception module preferentially checks whether or not the pair of received signals satisfies the equation 41. If the pair of received signals satisfies the equation 41, the reception module checks whether or not a phase difference of the pair of received signals satisfies equation 42 described in the following. α−δ≤λ_(k) ^(n)≤α+δ  [Equation 42]

where λ_(k) ^(n)=

(r_(k) ^(n+1)(r_(k) ^(n))*)

In the equation 42, α∈{0°, 90°, 180°, 270° } is satisfied and δ corresponds to a prescribed threshold range. For example, if the α corresponds to 90° and the δ corresponds to 10°, the equation 42 corresponds to 80°≤λ_(k) ^(n)≤100°. If the equation 42 is satisfied, a reception module is able to know that the pair of received signals corresponds to the (A-1), (B), and (C) cases and the pair of received signals can be used as a sample for estimating a residual CFO. Meanwhile, if the equation 42 is not satisfied, the reception module is able to know that the pair of received signals corresponds to the (A-2) case. In this case, the reception module does not use the pair of received signals for the CFO estimation procedure. Meanwhile, candidate values for the α are not restricted to 0°, 90°, 180°, and 270°. The candidate values can be configured by values of a different range adjacent to 0°, 90°, 180°, and 270°.

In case of the (A-2) case of FIG. 30, as mentioned in the foregoing description, a phase difference of the pair of received signals may correspond to one selected from the group consisting of {46°, 136°, 226°, 316°}. Hence, if the phase difference of the pair of received signals is calculated as being adjacent to a specific value, the pair of received signals can be used for estimating a CFO via a simple additional procedure.

According to an embodiment proposed in the present invention, if α∈{0°, 90°, 180°, 270°} and α∈{46°, 136°, 226°, 316°} are determined in the equation 42 and a phase difference λ_(k) ^(n) is determined as α∈{46°, 136°, 226°, 316°}, the reception module may change a phase of multiplication of the pair of received signals as much as

$e^{{- j}\frac{46{^\circ}}{180{^\circ}}\pi}$ according to equation 43 described in the following. Meanwhile, 46°, 136°, 226°, and 316° are just an example only. It may use different values adjacent to the 46°, 136°, 226°, and 316°.

$\begin{matrix} {y_{k}^{n} = {{r_{k}^{n + 1}\left( r_{k}^{n} \right)}^{*}e^{{- j}\frac{46{^\circ}}{180{^\circ}}\pi}}} & \left\lbrack {{Equation}\mspace{14mu} 43} \right\rbrack \end{matrix}$

According to the equation 43, a changed phase difference of the pair of received signals may correspond to one selected from the group consisting of 0°, 90°, 180°, and 270°. Hence, the phase difference changed by the equation 43 becomes similar to the (A-1) case and the pair of received signals can be used for estimating a CFO. In particular, if the phase of multiplication of the pair of received signals is changed, the number of pairs of received signals capable of being used for estimating a CFO increase. In particular, since a size of C increases, it may be able to enhance CFO estimation performance.

In the aforementioned embodiments, a reception module calculates a size ratio (or, a size difference) and a phase difference of a pair of received signals to determine whether or not the pair of received signal is used for estimating a CFO. Table 2 in the following illustrates that |C| varies according to a method for a reception module to manage an embodiment proposed in the present invention. Table 2 illustrates probability values that vary according to each of methods.

TABLE 2     Size ratio of pair of received signals Whether or not phase difference of pair of received signals is compensated (whether or not not phase changing embodiment is applied)   $p\left( \frac{\overset{\_}{C}}{C} \right)$ γ_(k = 1) ^(n) X 1/4 γ_(k = 1) ^(n) O 3/8 γ_(k = 1 or 3) ^(n) X 3/8 γ_(k = 1 or 3) ^(n) O 1/2

In Table 2, if γ_(k) ^(n) corresponds to 1 and a phase changing embodiment is not applied, (A-1), (b), and (C) of FIG. 30 are allowed only as samples for performing a CFO estimation method. In this case, a probability that two data symbols corresponding to a pair of received signals belong to (b) or (C) at the same time corresponds to (¼)*(¼)=( 1/16). A probability that two data symbols corresponding to a pair of received signals belong to (a) only corresponds to (½)*(½)=(¼). Since (A-1) is processed only, a probability for the (A-1) becomes (⅛). Consequently, a value of

$p\mspace{11mu}\left( \frac{\overset{\_}{C}}{C} \right)$ indicating a ratio of a pair of received signals, which are used for estimating a CFO, among the total reception signal pairs becomes ( 1/16)+( 1/16)+(⅛)=(¼).

Similarly, if γ_(k) ^(n) corresponds to 1 and a phase changing embodiment is applied, (A-1), (A-2), (b), and (C) are all allowed as samples for performing a CFO estimation method. In this case, a value of

$p\mspace{11mu}\left( \frac{\overset{\_}{C}}{C} \right)$ becomes ( 1/16)+( 1/16)+(¼)=(⅜).

If γ_(k) ^(n) corresponds to 1 or 3 and a phase changing embodiment is not applied, (A-1) of FIG. 30 and (b) of FIG. 31 are allowed as samples for performing a CFO estimation method. In this case, a probability that two data symbols corresponding to a pair of received signals belong to (b) of FIG. 31 at the same time corresponds to (½)*(½)=(¼). A probability that two data symbols corresponding to a pair of received signals belong to (A-1) corresponds to (⅛). Consequently, a value of

$p\mspace{11mu}\left( \frac{\overset{\_}{C}}{C} \right)$ becomes (¼)+(⅛)=(⅜).

If γ_(k) ^(n) corresponds to 1 or 3 and a phase changing embodiment is applied, (A-1) and (A-2) of FIG. 30 and (b) of FIG. 31 are allowed as samples for performing a CFO estimation method. Consequently, a value of

$p\mspace{11mu}\left( \frac{\overset{\_}{C}}{C} \right)$ becomes (¼)+(¼)=(½).

In particular, a reception module calculates a size ratio and a phase difference value of a pair of received signals and compares the calculated value with a predetermined number to determine whether or not the pair of received signals is used for a method of estimating a CFO. If more pairs of received signals are used for a CFO estimation method, CFO estimation performance can be increased. As mentioned in the foregoing description, the reception module may utilize the half of the total reception signal pairs according to 16-QAM for CFO estimation. Meanwhile, information on a maximum value for which a size ratio is allowed and information on whether or not a phase difference is compensated can be determined by a user in advance or can be systematically determined. For example, if channel environment is not sufficiently good, a reception module may allow a size ratio of 1 only to make a phase difference compensation not to be applied. In particular, the number of pairs of received signals utilized for a CFO estimation method can be adaptively changed according to communication environment.

In the foregoing description, an embodiment of similarly applying a blind type residual CFO estimation method, which was applied to BPSK, QBPSK, and QPSK, has been explained by segmenting 16-QAM. In this case, 16-QAM is just an example only for clarity. The proposed method can also be similarly applied to various modulation schemes such as 32, 64, 128, and 256 QAM.

FIG. 32 is a flowchart for a method of estimating a CFO according to the present invention. FIG. 32 illustrates the CFO estimation method according to the embodiments mentioned earlier in FIGS. 30 and 31 according to a time serial flow. Hence, although the embodiments mentioned earlier in FIGS. 30 and 31 are not explained in detail in FIG. 32, the embodiments can be similarly or identically applied.

First of all, a transmission module transmits data to a reception module [S3210]. The data can be transmitted in a unit of a frame defined by an OFDM symbol and a subcarrier. The data is transmitted to the reception module in a manner of being mapped to a specific constellation. The constellation may use various high order QAMs including 16, 32, 64, 128, and 256 QAM. The present embodiment is explained using an example of 16-QAM.

The reception module firstly estimates a CFO value from a received signal [S3220]. This procedure is performed using such a preamble part as L-STF, L-LTF, and the like of a frame. Yet, since the CFO value changes over time, the CFO value estimated in the step S3220 may not be an accurate CFO value. Hence, the reception module performs a procedure for estimating a residual CFO in succession.

First of all, the reception module selects a pair of received signals to be used for estimating a residual CFO value [S3230]. This procedure can be performed via a procedure for the reception module to calculate a size ratio of the pair of received signals. In addition to the procedure of calculating the size ratio of the pair of received signals, the reception module can also perform a procedure of calculating a phase difference of the pair of received signals. If the calculated size ratio and/or the phase difference corresponds to a specific value or a value belonging to a threshold range adjacent to the specific value, the reception module determines to use the pair of received signals for estimating a residual CFO. In particular, pairs of received signals are selected in the step S3230 as samples for estimating a residual CFO.

Subsequently, the reception module estimates a residual CF using the selected pairs of received signals [S3240]. This procedure can be performed via the second CFO estimation method mentioned earlier in FIGS. 24 to 26. In particular, the reception module interprets a part of a pair of received signals received through 16-QAM as QPSK and estimates a residual CFO using a residual CFO estimation method applied to a QPSK modulation scheme.

Lastly, the reception module eliminates a CFO estimated from a received signal and an impact of a residual CFO value to correctly decode the data transmitted by the transmission module.

7. Apparatus Configuration

FIG. 33 is a block diagram showing the configuration of a reception module and a transmission module according to one embodiment of the present invention. In FIG. 33, the reception module 100 and the transmission module 200 may include radio frequency (RF) units 110 and 210, processors 120 and 220 and memories 130 and 230, respectively. Although a 1:1 communication environment between the reception module 100 and the transmission module 200 is shown in FIG. 33, a communication environment may be established between a plurality of reception module and the transmission module. In addition, the transmission module 200 shown in FIG. 33 is applicable to a macro cell base station and a small cell base station.

The RF units 110 and 210 may include transmitters 112 and 212 and receivers 114 and 214, respectively. The transmitter 112 and the receiver 114 of the reception module 100 are configured to transmit and receive signals to and from the transmission module 200 and other reception modules and the processor 120 is functionally connected to the transmitter 112 and the receiver 114 to control a process of, at the transmitter 112 and the receiver 114, transmitting and receiving signals to and from other apparatuses. The processor 120 processes a signal to be transmitted, sends the processed signal to the transmitter 112 and processes a signal received by the receiver 114.

If necessary, the processor 120 may store information included in an exchanged message in the memory 130. By this structure, the reception module 100 may perform the methods of the various embodiments of the present invention.

The transmitter 212 and the receiver 214 of the transmission module 200 are configured to transmit and receive signals to and from another transmission module and reception modules and the processor 220 are functionally connected to the transmitter 212 and the receiver 214 to control a process of, at the transmitter 212 and the receiver 214, transmitting and receiving signals to and from other apparatuses. The processor 220 processes a signal to be transmitted, sends the processed signal to the transmitter 212 and processes a signal received by the receiver 214. If necessary, the processor 220 may store information included in an exchanged message in the memory 230. By this structure, the transmission module 200 may perform the methods of the various embodiments of the present invention.

The processors 120 and 220 of the reception module 100 and the transmission module 200 instruct (for example, control, adjust, or manage) the operations of the reception module 100 and the transmission module 200, respectively. The processors 120 and 220 may be connected to the memories 130 and 230 for storing program code and data, respectively. The memories 130 and 230 are respectively connected to the processors 120 and 220 so as to store operating systems, applications and general files.

The processors 120 and 220 of the present invention may be called controllers, microcontrollers, microprocessors, microcomputers, etc. The processors 120 and 220 may be implemented by hardware, firmware, software, or a combination thereof.

If the embodiments of the present invention are implemented by hardware, Application Specific Integrated Circuits (ASICs), Digital Signal Processors (DSPs), Digital Signal Processing Devices (DSPDs), Programmable Logic Devices (PLDs), Field Programmable Gate Arrays (FPGAs), etc. may be included in the processors 120 and 220.

Meanwhile, the aforementioned method may be implemented as programs executable in computers and executed in general computers that operate the programs using computer readable media. In addition, data used in the aforementioned method may be recorded in computer readable recording media through various means. It should be understood that program storage devices that can be used to describe storage devices including computer code executable to perform various methods of the present invention do not include temporary objects such as carrier waves or signals. The computer readable media include storage media such as magnetic recording media (e.g. ROM, floppy disk and hard disk) and optical reading media (e.g. CD-ROM and DVD).

It will be apparent to those skilled in the art that various modifications and variations can be made in the present invention without departing from the spirit or scope of the inventions. Thus, it is intended that the present invention covers the modifications and variations of this invention provided they come within the scope of the appended claims and their equivalents.

INDUSTRIAL APPLICABILITY

Although the CFO estimation method is mainly described based on an IEEE 802.11 system and an HEW system, the method can be applied to various kinds of wireless communication systems as well as the IEEE 802.11 system and the HEW system. 

What is claimed is:
 1. A method of estimating a carrier frequency offset (CFO) by a reception module in a wireless communication system using 16-quadrature amplitude modulation (QAM), the method comprising: measuring a size ratio between a pair of signals received on two consecutive orthogonal frequency division multiplexing (OFDM) symbols for a specific subcarrier; based on determination that the size ratio belongs to a first threshold range, determining the pair of received signals as a pair to be used for estimating a residual CFO; and estimating the residual CFO using only the pair of received signals that are determined to be used for estimating the residual CFO.
 2. The method of claim 1, further comprising measuring a phase difference between the pair of received signals, wherein when the size ratio belongs to the first threshold range and the phase difference belongs to a second threshold range, the pair of received signals is determined as a pair to be used for estimating the residual CFO.
 3. The method of claim 2, further comprising when the phase difference does not belong to the second threshold range and belongs to a third threshold range, changing a phase of multiplication of the pair of received signals as much as a predetermined value.
 4. The method of claim 3, further comprising when the phase difference does not belong to the second threshold range and does not belong to the third threshold range, determining not to use the pair of received signals for estimating the residual CFO.
 5. The method of claim 3, wherein the second threshold range indicates a range within a predetermined value from one value among 0°, 90°, 180°, and 270°.
 6. The method of claim 3, wherein the third threshold range indicates a range within a predetermined value from one value among 46°, 136°, 226°, and 316°.
 7. The method of claim 1, wherein the first threshold range indicates a range within a predetermined value from one value among 1, ⅓, and
 3. 8. The method of claim 1, wherein the step of estimating the residual CFO estimates the residual CFO according to a blind scheme which is applied when quadrature phase shift keying (QPSK) is used.
 9. A communication device for estimating a carrier frequency offset (CFO) in a wireless communication system using 16-quadrature amplitude modulation (QAM), the communication device comprising: a memory; and a processor operably coupled with the memory and configured to: measure a size ratio between a pair of signals received on two consecutive orthogonal frequency division multiplexing (OFDM) symbols for a specific subcarrier; based on determination that the size ratio belongs to a first threshold range, determines the pair of received signals as a pair to be used for estimating a residual CFO; and estimate the residual CFO using only the pair of received signals that are determined to be used for estimating the residual CFO.
 10. The communication device of claim 9, wherein the processor is further configured to: measure a phase difference between the pair of received signals; and when the size ratio belongs to the first threshold range and the phase difference belongs to a second threshold range, determine the pair of received signals as a pair to be used for estimating the residual CFO.
 11. The communication device of claim 10, wherein the processor is further configured to, when the phase difference does not belong to the second threshold range and belongs to a third threshold range, change a phase of multiplication of the pair of received signals as much as a predetermined value.
 12. The communication device of claim 11, wherein the processor is further configured to, when the phase difference does not belong to the second threshold range and does not belong to the third threshold range, determine not to use the pair of received signals for estimating the residual CFO.
 13. The communication device of claim 11, wherein the second threshold range indicates a range within a predetermined value from one value among 0°, 90°, 180°, and 270°.
 14. The communication device of claim 11, wherein the third threshold range indicates a range within a predetermined value from one value among 46°, 136°, 226°, and 316°.
 15. The communication device of claim 9, wherein the first threshold range indicates a range within a predetermined value from one value among 1, ⅓, and
 3. 16. The communication device of claim 9, wherein the processor is further configured to estimate the residual CFO according to a blind scheme which is applied when quadrature phase shift keying (QPSK) is used. 